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content/wire/160new.html

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+<!DOCTYPE html>
+<html lang="en">
+<head>
+  <meta charset="UTF-8">
+  <meta name="keywords" content="MF, Vertical, Inverted L, Dipole, Horizontal Loop">
+  <title>Your First 160-Meter Antenna</title>
+  <link rel="stylesheet" href="../styles.css">
+</head>
+<body>
+  <center>
+    <h1><img src="../images/wirecad.gif" align="left" width="125" height="62" border="2" vspace="15" hspace="9"> <img src="../images/w4rnlwwr.gif" align="right" width="99" height="83" border="2" vspace="5" hspace="10"> Your First 160-Meter Antenna</h1>
+  </center>
+  <center>
+    <h3>L. B. Cebik, W4RNL</h3>
+  </center>
+  <center>
+    <img src="../images/colorbar.gif" align="middle" width="540" height="4">
+  </center>
+  <p>The following notes rest on a small set of assumptions.</p>
+  <blockquote>
+    <p>1. You want to get on 160 meters for the first time (or perhaps, for the first time in a long time).</p>
+    <p>2. You want to set up the simplest possible effective antenna using all wire construction. In fact, all of the antennas will be made from AWG #14 or AWG #12 wire. 2-mm (0.0787") diameter wire falls right between these sizes, so all of the data will use that value. However, nothing much changes by reducing the diameter to AWG #14 (0.0641") or increasing the diameter to AWG #12 (0.0808").</p>
+    <p>3. You do not have unlimited vertical space for your antenna. In these notes, the limit will be about 70'. In fact, I shall use 21 m (68.9') as the standard top height for all antennas.</p>
+  </blockquote>
+  <p>I have set these limits so that we can compare the performance of a collection of relatively simple antennas.</p>
+  <p>For all comparisons, we shall use average ground with a conductivity of 0.005 S/m and a relative permittivity (dielectric constant) of 13. For vertical antennas especially, you should expect lesser performance from worse ground and better performance from better ground--but not radically worse or better. Horizontal antennas are less affected by ground quality, but the top height is so low (about 1/8 wavelength) that the ground will influence performance much more than for the antennas you place 1 wavelength above ground for the upper HF region.</p>
+  <p>160-meter antennas are naturally much larger (longer, taller) than antennas for the other HF amateur bands. Therefore, be prepared to spend a little more money for quality wire and insulators to durably bear the antenna weight. Copperweld is desirable. Supporting structures--whether natural or constructed--need to be stronger and taller than the average sorts of things that populate a backyard. How you handle the support structures I shall leave to you, since every yard is different, as are the locally available materials and the construction skills at hand.</p>
+  <p>With those qualifications, let's get started in our work, starting with some vertical antennas.</p>
+  <p><b>160-Meter Wire Verticals</b></p>
+  <p>We shall begin with an antenna that violates the upper height limit of our task: the full-size 1/4 wavelength vertical monopole. A wire version of this antenna needs to be about 39 m (128') tall. The convenience of the vertical monopole is that we can feed it at the base--at or near ground level. The inconvenience is that we must install radials. The radials should be about 1/4 wavelength long and placed as symmetrically as the yard space allows. To see how many radials we might need, I modeled the vertical using 4, 16, and 64 1/4 wavelength radials, each 6" (0.15 m) below the surface. <b>Fig. 1</b> shows the outlines of the 3 models.</p>
+  <center>
+    <img src="160new-1.gif" width="660" height="190" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The following table shows the anticipated results, assuming that the vicinity of the antenna is not filled with RF-eating ground clutter. Conductive objects--even semi-conducting trees and shrubs--can distort antenna patterns and absorb some RF energy, so keeping the antenna area as clean as possible is important to getting the most out of any vertical antenna.</p>
+  <pre>
+1/4-Wavelength Vertical Monopole with Variable Radial Systems
+Average Ground
+No. of            Maximum          TO Angle         Feedpoint Z
+Radials           Gain dBi         degrees          R +/- jX Ohms
+ 4                -0.72            23               57 + j 1
+16                0.48             23               44 - j 8
+64                1.14             23               37 - j12
+</pre>
+  <p>Note that we gain about 1.2 dB by increasing the radial field from 4 to 16 wires, with another increase of about 0.7 dB by raising the count to 64. <b>Fig. 2</b> shows the relative radiation pattern strengths. The radiation plot also shows that a vertical antenna is best for lower-angle long-distance skip signals, but almost unusable for NVIS (Near Vertical Incidence Skywave) very short distance communications. Many vertical users also find a vertical less noisy that a horizontal antenna in terms of QRN from lightning, but more susceptible to local man-made noise sources. As well, as we increase the number of radials, the impedance decreases, indicating a reduction in energy lost to the ground.</p>
+  <center>
+    <img src="160new-2.gif" width="426" height="415" border="2" vspace="5" hspace="9">
+  </center>
+  <p>As the impedance decreases due either to the number of radials or ground quality, a number of operators use a simple means of obtaining a good match for coaxial cable. By making the vertical longer, they increase the resistive component of the impedance and the reactance moves from being slightly capacitive to being more definitely inductive. Adding a series capacitor at the feedpoint between the cable center conductor and the feedpoint itself allows them to compensate for the reactance, leaving a nearly perfect match for the 50-Ohm cable. A fixed capacitor may work if you have a specific operating frequency, but a remotely tuned variable is necessary for obtaining a low SWR over a wider operating bandwidth. Since we want the antenna to be at least slightly inductively reactive at all operating frequencies, setting up the antenna for the low edge of the band is the usual practice.</p>
+  <p>The full-size vertical monopole is useful as a reference for comparing other vertically polarized antenna candidates. With that data, we can see what we gain or lose from each one. We shall look at 2 candidates, each no more than 70' tall.</p>
+  <p><i>The Tee-Vertical</i>: If we must limit the height to a certain level--70' in our case--but still desire a perfectly circular pattern, we need to create a shorter vertical antenna. Many vertical users opt for inductively loading the vertical either at its base or higher up on the wire. However, inductive loading has two disadvantages. First, the inductor always has a series resistance that reduces the radiated energy. Second, inductive loading reduces the feedpoint impedance faster, the closer the inductor is to the feedpoint.</p>
+  <p>One of the simplest and most efficient ways to shorten a vertical monopole is to create a hat at the top. The usual vision of a hat consists of several hat wires radiating from the top of the vertical wire. However, we actually need only 2 wires to effect a hat. (The more wires that we have in a symmetrical arrangement, the shorter that each must be to set the antenna at resonance. However, any wires not in the same line as the supports for the top of the vertical section require additional supports.)</p>
+  <p>We shall look at 3 versions of a Tee-vertical: with 4, 16, and 64 radials. <b>Fig. 3</b> shows the relative complexity of each version. The vertical wire is 21 m (68.9'), and each leg of the Tee is 11.6 m (38') long.</p>
+  <center>
+    <img src="160new-3.gif" width="660" height="160" border="2" vspace="5" hspace="9">
+  </center>
+  <p>For the same set of conditions used to model the full-size vertical monopole, the shortened Tee-vertical shows the following performance values.</p>
+  <pre>
+Shortened Tee-Vertical Monopole with Variable Radial Systems
+Average Ground
+No. of            Maximum          TO Angle         Feedpoint Z
+Radials           Gain dBi         degrees          R +/- jX Ohms
+ 4                -1.45            25               42 + j 2
+16                0.20             25               29 - j 6
+64                1.11             25               23 - j11
+</pre>
+  <p>The fewer the radials, the more that Tee-vertical performance lags behind the performance of the full-size vertical monopole. With 64 radials, there is almost no difference in performance with respect to gain. The Shorter vertical section of the Tee version does show a 2-degree increase in the TO angle. As well, the impedance at the feedpoint is only about 70% of the value for the full-size vertical. <b>Fig. 4</b> shows the relative radiation patterns. We do not need azimuth patterns because, like the full size vertical, the Tee-vertical provides virtually a perfect circle of radiation (assuming that there are no nearby objects to distort that pattern).</p>
+  <center>
+    <img src="160new-4.gif" width="426" height="415" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The Tee-vertical is amenable to the use of lengthening techniques to raise the feedpoint impedance with a series capacitor to compensate for the inductive reactance. Lengthening the Tee legs (in equal amounts to preserve symmetry) saves you the trouble of increasing the height. However, you will need more horizontal space for the increased Tee-top. If you use a series capacitor at the base of the antenna, I recommend a double waterproofing case system, along with regular preventive maintenance. As well, be sure that you use a beefy capacitor able to handle the high current level at a 50-Ohm impedance.</p>
+  <p><i>The 1/4-Wavelength Inverted-L</i>: A second alternative for our 70' height limitation is the inverted-L. As shown in <b>Fig. 5</b>, the L does not worry about symmetry, but simply uses a horizontal extension of the vertical wire to reach resonance on 160 meters. Because the top is not symmetrical, the horizontal wire radiates. However, the current is lower in the horizontal part of the antenna and the pattern is not seriously distorted on 160 meters. In the model for 1.85 MHz over average ground, the horizontal wire is 19 m (62.3') for the same vertical wire used in the Tee-vertical.</p>
+  <center>
+    <img src="160new-5.gif" width="660" height="160" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The performance of the inverted-L is not significantly different from the Tee, as shown by the following performance figures.</p>
+  <pre>
+Shortened Tee-Vertical Monopole with Variable Radial Systems
+Average Ground
+No. of            Maximum          TO Angle         Feedpoint Z
+Radials           Gain dBi         degrees          R +/- jX Ohms
+ 4                -1.53            26               43 + j 3
+16                0.08             26               30 - j 6
+64                0.98             26               24 - j11
+</pre>
+  <p>Due to the small horizontal component of the radiation patterns, the elevation angle has increase by another degree. However, the impedance values are almost identical to the corresponding values for the Tee-vertical. <b>Fig. 6</b> shows the elevation and the azimuth patterns for the inverted-L. Note that the presence of a non-symmetrical horizontal section does not allow the pattern overhead to go to nearly zero, although the level is not strong enough for effective NVIS communications. The azimuth pattern shows a slight push in the direction of the top section of the L. However, the differential is not large enough to be noticed during operation.</p>
+  <center>
+    <img src="160new-6.gif" width="800" height="415" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The 160-m 1/4 wavelength inverted-L has another advantage. With a wide-range tuner at the feedpoint (perhaps one of the remote tuners on today's market), the antenna is usable for general communications on virtually all of the amateur bands. Above 160-meters, the radial system acts like a good RF ground between the operating position and the antenna base, since the antenna is 1/2 wavelength or longer on all bands above 160 meters. If you choose to use a remote tuner for such an inverted-L system, add another layer of water-proofing as an additional guard against weather penetration of the tuner and the connection. For further information on multi-band use of the inverted-L, see "<a href="../fdim/fdim10.pdf">Straightening Out the Inverted-L</a>."</p>
+  <p>There is one temptation to avoid with the 160-meter 1/4 wavelength inverted-L. Many operators obtain rather poor results because they place the vertical section of the antenna too close to a natural or man-made support. The vertical section needs as much clearance from other objects as the corresponding part of the full-size and the Tee verticals.</p>
+  <p><b>160-Meter Wire Horizontal Antennas</b></p>
+  <p>We have examined the main candidates for vertical wire antennas, although there are manmy variations on the basic designs that we have used as examples. We should also look at some horizontal basic wire antennas. Any horizontal antenna will be severely limited by the 70' height restriction that we placed on the exercise. 70' is only about 1/8 wavelength above ground, a height that is even below optimum for NVIS operation--although it will work quite well in this service. One advantage of the horizontal wire is that it does not require any radials. A second advantage--at least for our work--is that horizontal wires do not change performance characteristics very much as we change ground quality. Therefore, the use of average ground provides a good indication of operation over any soil type. Finally, there are only 2 important horizontal variations that are possible within our height restriction: linear wires and closed horizontal loops.</p>
+  <p><i>The 1/2-Wavelength Dipole</i>: There is no magic about the 1/2 wavelength dipole except that at resonance, it is a reasonably good match for coaxial cable. If we wish to use parallel feedline and a tuner, we can be less critical about the exact length without changing the pattern in any detectable way. <b>Fig. 7</b> shows the details of our model set-up. The wire is 78 m (256') long.</p>
+  <center>
+    <img src="160new-7.gif" width="480" height="150" border="2" vspace="5" hspace="9">
+  </center>
+  <p>Since we have only a single model with which to deal, our performance table is simplified.</p>
+  <pre>
+1/2-Wavelength Horizontal Dipole 70' above Average Ground
+    Maximum          TO Angle         Feedpoint Z
+    Gain dBi         degrees          R +/- jX Ohms
+    6.72             90               49 + j 0
+</pre>
+  <p>Note that the horizontal wire provides the strongest radiation (and receiving sensitivity) straight up. <b>Fig. 8</b> compares the elevation pattern of the dipole with the elevation pattern for the inverted-L with 16 radials. The horizontal wire is superior for NVIS service, but inferior for long-range, low-angle service. The horizontal wire is likely to be more susceptible to lightning noise, but less susceptible to man-made noises. The patterns for the two types of antennas cross at about the 23-degree elevation mark.</p>
+  <center>
+    <img src="160new-8.gif" width="426" height="415" border="2" vspace="5" hspace="9">
+  </center>
+  <p>Despite the low height of the dipole when registered as a fraction of a wavelength, the azimuth pattern at almost any elevation angle is still bi-directional and broadside to the wire. <b>Fig. 9</b> shows the azimuth pattern at a lower angle (25 degrees elevation). Radiation (and reception) off the ends of the wire is about 8-dB or about 1.5 S-units weaker than broadside to the wire.</p>
+  <center>
+    <img src="160new-9.gif" width="426" height="415" border="2" vspace="5" hspace="9">
+  </center>
+  <p>Linear wires with open ends can build considerable levels of static charge unless we take measures to bleed it off as it develops. One technique is to place either a high-value resistor or an RF choke across the antenna feedpoint, ensuring that one side is connected to the coax braid--and the coax braid is well grounded. Inserting a transmission-line transformer type of balun at the feedpoint will defeat this measure by physically isolating the feedpoints from the cable braid. However, using a W2DU-type ferrite-bead choke as the balun will allow the bleed-off component to do its work.</p>
+  <p><i>The 2-Wavelength Horizontal Loop</i>: A closed loop antenna is more immune to static charge build-up, but has some special requirements. To understand why the heading specifies a 2 wavelength circumference for the loop horizontal antenna, we should proceed a step at a time. Let's begin with a simple square loop, like the one shown in <b>Fig. 10</b>. Our initial exercise will place the loop in free space and vary the circumference from 1.0 to 2.5 wavelengths.</p>
+  <center>
+    <img src="160new-10.gif" width="290" height="250" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The following table lists the free-space performance values for the loop. The column marked "Horizontal Gain" lists the gain in the plane of the loop. The column labeled "Vertical Gain" shows the gain broadside to the face of the loop.</p>
+  <pre>
+Free-Space Performance of Horizontal Loops of Various Sizes
+Circumference     Horizontal     Vertical       Feedpoint Z
+WL                Gain dBi       Gain dBi       R +/- jX Ohms
+1.0               0.09           3.27            124 + j   17
+1.5               1.49           2.97           5300 - j 4700
+2.0               3.07           0.18            300 + j  240
+2.5               2.06           1.09           2600 - j 2700
+</pre>
+  <p>The 1 wavelength loop is most useful in parasitic beams called quads, where the individual loops are set up vertically to take advantage of the stronger radiation broadside to the plane of the loop. However, when we place the loop horizontally over ground, the radiation from the edge of the loop--the plane of most interest--is much weaker. As the table shows, the edge, in-plane, or "horizontal" radiation is strongest when the loop is about 2 wavelengths in circumference. For our test model, that length is about 340 m (1115'). Since the loop is not resonant, we shall need parallel transmission line and a tuner. Hence, the exact length is not at all critical. Any total circumference around 1100' will work fine.</p>
+  <center>
+    <img src="160new-11.gif" width="800" height="415" border="2" vspace="5" hspace="9">
+  </center>
+  <p><b>Fig. 11</b> compares the elevation and azimuth patterns for 1 and 2 wavelength loops 70' above average ground. Note that due to the low height, even the 2 wavelength loop has a relatively high TO angle. However, the 2 wavelength radiation strength (and reception sensitivity) at lower angles is considerably greater than the 1 wavelength loop. The advantage at lower angles appears clearly in the azimuth patterns on the right. The "tilt of the pattern follows the placement of the feedpoint, shown in <b>Fig. 10</b>. Note that the 2 wavelength loop does not produce a circular--or even an oval--pattern. Rather, it has four wide major lobes. The following table completes the equivalent data for all of the loop sizes that we tested in free-space. Note that the impedance reports change relative to the free-space values--as a function of the low height of the antennas above ground. The resistive component is lower, while the reactive component is more inductive.</p>
+  <pre>
+Performance of Horizontal Loops of Various Sizes 70' above Average Ground
+Circumference     Maximum        TO Angle       Feedpoint Z
+WL                Gain dBi       degrees        R +/- jX Ohms
+1.0               7.38           90              100 + j  100
+1.5               6.65           90             2600 - j 5200
+2.0               5.65           50              200 + j  380
+2.5               6.02           53             1400 - j 3300
+</pre>
+  <p>The pattern shapes and TO angles for a horizontal loop change as we change the shape of the loop. They also change if we move the feedpoint, say, from a corner to the middle of a side. As samples of the sort of changes that we might encounter with relatively symmetrical simple structures, I modeled triangular, square, and hexagonal loops, feeding each structure both at a corner and in the middle of a side. The following table summarizes the results. It adds a column listing the maximum gain at a "standard" 30-degree elevation angle, since the TO angle is considerably higher in most cases and varies from case to case.</p>
+  <pre>
+Performance of 2-Wavelength Horizontal 70' above Average Ground
+Loop and          Maximum        TO Angle       Gain at 30-deg       Feedpoint Z
+Feed positionL    Gain dBi       degrees        dBi                  R +/- jX Ohms
+Triangle-Corner   6.05           54             3.56                 135 + j 315
+Triangle-Side     5.99           58             3.18                 225 + j 300
+Square-Corner     4.92           55             1.24                  75 + j 220
+Square-Side       5.65           50             3.51                 200 + j 380
+Hexagon-Corner    5.65           53             2.75                 140 + j 320
+Hexagob-Side      5.57           54             2.45                 145 + j 320
+</pre>
+  <p>The wires of a 2 wavelength loop interact with each other to produce distinctive patterns for each combination of overall shape and feedpoint placement. <b>Fig. 12</b> shows the azimuth patterns for the two triangles, with plots taken at the TO angle at at a standard 30-degree elevation angle. The insets show the loop outline and the feedpoint placement relative to the pattern for each version of the triangle. In all of the plots of 2 wavelength horizontal loops, the feedpoint will be at the top or 0-degree azimuth direction.</p>
+  <center>
+    <img src="160new-12.gif" width="800" height="415" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The two triangle patterns are similar, although there is a small displacement of the pattern toward the long-wire side and away from the triangle point. More significant is the fact that in both cases, the pattern is significantly stronger (by about 3 dB) along a line from the feedpoint through the center than from side to side. Otherwise, there is not much to choose between the two versions of the triangle.</p>
+  <center>
+    <img src="160new-13.gif" width="800" height="415" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The patterns in <b>Fig. 13</b> confirm what the data in the table suggest: the feedpoint position makes a much more important difference to performance with a square loop than with any other form. With a corner feed, we obtain nearly circular patterns, but at lower strength. With a side-feed, we obtain more gain, but the patterns take on the 4-lobe shape. The lower the elevation angle, the more distinct that the lobes become. Whether the pattern shape and gain provide an advantage may depend on the possibilities for laying out the antenna relative to desired communication targets.</p>
+  <center>
+    <img src="160new-14.gif" width="800" height="415" border="2" vspace="5" hspace="9">
+  </center>
+  <p>As we make the loop more circular, the exact shape and feedpoint make less difference to performance. The hexagon patterns appear in <b>Fig. 14</b>. Neither the pattern shape nor the gain change very much as we re-orient the loop and the feedpoint. As well, the corner-fed and side-fed versions of the loop exhibit feedpoint impedance values that are much closer together than for either the triangle or the square.</p>
+  <p>The most desirable version of a 2 wavelength horizontal loop would be a circle. However, the realities of antenna construction will not only require simpler forms, but as well, they may dictate somewhat irregular shapes. Nonetheless, virtually any horizontal loop will provide very reasonable performance. In addition, unlike a dipole, they will provide a null overhead, much like the nulls of vertical antennas. Therefore, if NVIS operation is the goal, you much either create a 1 wavelength loop or a dipole. For operation in the 20-30-degree elevation range, the 2 wavelength loop will usually provide as much or more gain than a wire vertical. <b>Fig. 15</b> compares the elevation patterns of the corner-fed hex loop and the inverted-L with 16 radials. The maximum gain limits of the loop are similar to those of the dipole at the same 70' height, but the pattern is nearly circular rather than being bi-directional.</p>
+  <center>
+    <img src="160new-15.gif" width="426" height="415" border="2" vspace="5" hspace="9">
+  </center>
+  <p>Both the 160-meter dipole and the 2 wavelength loop are useful as multi-band antennas if we feed them with parallel transmission line and employ an antenna tuner to achieve a match with the transceiver. A number of other items at this site address the kinds of patterns that we can expect from a 250+' doublet and from horizontal loops (HOHPLs) of various shapes across the HF region.</p>
+  <p><b>Conclusion</b></p>
+  <p>We have surveyed some of the simplest antennas used on 160 meters. They are simple in principle, but require a lot of wire, whether used in the element or in radials. Insulation on the wire makes virtually no difference to performance. As noted early on, element wire should be strong, and copperweld is desirable. However, radials may use virtually any wire available. If a sale on wire allows you to add more radials to a vertical system, then it is worth the price. However, exposed elements require good strength or additional supports. As well, use good non-conductive insulators wherever an elevated wire terminates or changes direction. Do not lay a wire directly over a tree limb or wood support. High voltage has been known to gradually sever limbs or to set dry limbs ablaze. Suspend an insulator below the support and run the wire through the insulator. Likewise, use a strain relief fixture for any connection between the element and parallel transmission line.</p>
+  <p>We have not examined a number of excellent antenna systems, such as phased or parasitic verticals. 160-meter wire Yagis and LPDAs are also possible. These are advanced projects, and our mission was to set out and compare some basic antennas. However, eventually, you will wish to purchase a copy of ON4UN's book on <i>Low-Band DXing</i>. It is possibly the best collection of 160-meter (and 80- and 40-meter) antenna ideas available.</p>
+  <center>
+    <img src="../images/colorbar.gif" width="540" height="4">
+  </center>
+  <p><i>Updated 6-7-2005. &COPY; L. B. Cebik, W4RNL. Data may be used for personal purposes, but may not be reproduced for publication in print or any other medium without permission of the author.</i></p>
+  <center>
+    <img src="../images/colorbar.gif" width="540" height="4">
+  </center>
+  <p><a href="../radio.html"><img src="../images/rtnhome.gif" align="bottom" width="40" height="40"> Return to Index</a></p>
+  <p class="topics-footer">Topics: <a href='../topics.html#dipole'>Dipole</a>, <a href='../topics.html#horizontal-loop'>Horizontal Loop</a>, <a href='../topics.html#inverted-l'>Inverted L</a>, <a href='../topics.html#mf'>MF</a>, <a href='../topics.html#vertical'>Vertical</a></p><br>
+</body>
+</html>

content/wire/1712.html

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+<!DOCTYPE html>
+<html lang="en">
+<head>
+  <meta charset="UTF-8">
+  <meta name="keywords" content="HF Upper, Dipole, Doublet">
+  <title>17-15-12 and Simple</title>
+  <link rel="stylesheet" href="../styles.css">
+</head>
+<body>
+  <center>
+    <h1><img src="../images/wirecad.gif" align="middle" width="125" height="62" border="2" vspace="5" hspace="9"> 17-15-12 and Simple <img src="../images/w4rnlwwr.gif" align="middle" width="99" height="83" border="2" vspace="5" hspace="10"></h1>
+  </center>
+  <center>
+    <h3>L. B. Cebik, W4RNL</h3>
+  </center>
+  <center>
+    <img src="../images/colorbar.gif" align="middle" width="540" height="4">
+  </center><br>
+  <br>
+  <p>A number of casual CW operators have expressed interest in the 17-, 15-, and 12-meter bands as favorites for operating. QRM is less, in general, and the "atmosphere" seems relatively friendly. The smaller population on at least the 2 WARC bands in this collection also makes them easier for the less experienced operator to use.</p>
+  <p>A common thread of conversations about antennas for these bands runs something like this: The lobes of the 135' doublet seem to prevent full coverage of all directions. Even though it has somewhat more gain near the ends, the lobes are narrow with deep nulls between.</p>
+  <center>
+    <img src="1712-1.gif" width="640" height="480" border="2" vspace="5" hspace="9">
+  </center>
+  <p>As <b>Fig. 1</b> demonstrates, these problems are correctly diagnosed. Compared to dipole cut for 17 meters, the pattern for the doublet on 17 has many thin fingers. On 15 and 12, there will be more fingers, and thinner.</p>
+  <p>However, the typical antenna described as being just for these bands is usually a complex beam of interlaced elements. Construction is usually tricky. A commercial version is expensive. So the conversation next poses this question: Is there something simpler, less expensive, and fuller in beamwidth coverage that might be used?</p>
+  <p>The answer is yes. Unfortunately, the idea of using drooping dipole fans turns out not to be the most desirable option. The closeness of frequency of the 3 bands makes the interaction quite strong, even with sizable angles between wires. Hence, feeding one antenna array with a single coax cable may not be the easiest route to success. It can be done with careful pruning, but there may be an easier way.</p>
+  <p>We can use a fixed or a rotatable dipole for 17 meters fed with parallel transmission line to an antenna tuner (the one used with the doublet). By cutting the dipole for 17 meters, the antenna will still be only about 2/3 wl long at 12 meters. Hence, the pattern will not change its shape from the dipole figure-8 (in free space) by enough to matter as we raise the frequency through 21 and stop at 24.99 MHz.</p>
+  <center>
+    <img src="1712-2.gif" width="640" height="480" border="2" vspace="5" hspace="9">
+  </center>
+  <p><b>Fig. 2</b> compares the patterns for a 17-meter dipole in free space in the three desired bands. Operationally, the differences in patterns would not be detectable.</p>
+  <p>One important feature will change: the feedpoint impedance. If resonated in the middle of 17 meters, the impedance will be about 72 Ohms. On 15 meters, this same antenna will show a feedpoint impedance of about 130 + j200 Ohms. On 12 meters, the impedance will be in the neighborhood of 270 + j500 Ohms. The exact reactance value will vary with the wire size. 50-Ohm SWR values can get as high as 17:1 to 20:1.</p>
+  <p>The upshot is this: if the antenna is to be used on all three bands, it needs to be fed with parallel transmission line and an antenna tuner. This move also frees us from having to use ultra-precision in adjusting the length of the dipole.</p>
+  <p>If the antenna is placed at least 1/2 wl up in the air, skip performance will be excellent broadside to the antenna. A half wavelength at 17 meters is only about 27' up, so anything above that level is a bonus in terms of lowering the elevation angle of the main radiation lobes. The beamwidth of the dipole pattern is wide enough to make pointing very non-critical.</p>
+  <p>A thirty-foot TV mast might be pressed into duty for this antenna. The metal mast will require some attention to dressing the parallel feedline down to the shack to avoid unbalancing influences, but most handbooks have sage advice in this department. The mast might be set up to turn with a rotator or by hand--what used to be called the Armstrong method. A short piece of strong tubing mounted to the mast with a plate and U-bolts would make room for a removable rod or thinner pipe to ease the turning. The system described here is designed to let you store the turning pipe when not in use for family safety. Other methods both simple and complex can be used.</p>
+  <p>The system just described calls for an antenna that can be supported at the center at the mast top. See <b>Fig. 3</b>.</p>
+  <center>
+    <img src="1712-3.gif" width="627" height="274" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The sketch shows two ways (among many) to make up the dipole. The tubing sketch makes use of standard aluminum tubing available from places like Texas Towers and others. It comes in 6' lengths and the sizes nest closely in each other. Hence, little work is necessary to lock smaller sections inside larger ones. Handbooks, show hose clamp, bolt, sheet metal screw, and rivet methods of element joining. Pick the one most easily implemented, but do not skimp on the effort to make the junctions secure.</p>
+  <p>The tubing lengths in the diagram have a plan. The 33" exposed sections are all 36" long, with a 3" insertion length into the next larger tube section. This feature allows the builder to use a single 6" length of each size--and 6' is the standard shipping length. The 1" sections need 2 6' sections, with a bit left over. (A short section of fiberglass rod or tubing at the center of the 1" sections and running at least the length of the plywood plate can keep the elements aligned and strengthen the mounting.) Hence, even with shipping charges, the materials for this rotatable version of the 3-band dipole are inexpensive.</p>
+  <p>The center plate can be a 1' to 1.5' per edge square of exterior grade plywood (exterior on both sides). Orient the plate as a diamond. Several coats of spar varnish weather protect the plate. It pays to use an exterior grade fill on the plywood edges before varnishing. U-bolts hold the element to the plate (4: 2 for each side) and the plate to the mast (1 U-bolt top and bottom). There are sources, like Harbach, for U-bolt assemblies designed specifically for antenna work. Hardware depot supplies should be carefully checked to ensure they are stainless steel.</p>
+  <p>Some of the possible construction ideas appear in <b>Fig. 4</b>.</p>
+  <center>
+    <img src="1712-4.gif" width="658" height="435" border="2" vspace="5" hspace="9">
+  </center>
+  <p>Use any kind of insulated stand-offs to space the parallel transmission line from the mast. Then get plenty of help to raise things into position, thinking safety all the while. A 26.5' antenna can be somewhat ungainly.</p>
+  <p>The fixed wire version can be used, although getting it to rotate is more difficult. However, bamboo may still be available in some areas. Otherwise, one has to live with a fixed wire, so be sure the antenna is broadside to your favorite parts of the world.</p>
+  <p>Notice that the thinner wire dipole is actually shorter than the aluminum tapered diameter element. The particular set of element diameters and section lengths used in the tubing model actually is equivalent to a uniform diameter tube about 0.8" in diameter and 310" long. Tapering element diameters downward towards the outer ends calls for longer physical lengths longer than uniform diameter tubes.</p>
+  <p>The actual length of the wire version will depend on the wire used. Insulated wire usually requires some physical shortening, since insulation gives antenna wire a velocity factor between .95 and .99 depending on the insulating material and its thickness. However, if we use the parallel feedline, length is less critical since we are no longer striving for a perfect impedance match at any frequency of use.</p>
+  <p>Although designed for 17-12 meters, the antenna can be useful up to 10 meters and down to 30 meters. At these frequencies, the tuner might have some difficulty in reaching a match, but changing the parallel line length slightly may present the tuner with values of reactance it can more easily handle.</p>
+  <p>A rotatable dipole can be a very effective antenna, since it eliminates QRM off the sides. (A beam reduces QRM on only one more side than the dipole.) Its broad beamwidth makes orientation easy. On many days, it is only necessary to reorient the antenna in late morning and again in late afternoon. (Of course, on other days, skip may come from anywhere, and the run to the antenna for a change of direction can be counted as good exercise.) The simplicity of construction, inexpensive materials, and ease of maintenance go a long way to compensate for lower gain relative to complex, expensive Yagis. Parallel feedline keeps losses low and places the matching problem next to the rig, where it can be handled with ease.</p>
+  <p>If you already have your towers and beams, by all means use them. But for the casual CW operator who enjoys a rag chew on the upper bands or the pursuit of QRP with less QRM, a rotatable dipole for 17-15-12 meters may be just the ticket to getting started.</p>
+  <center>
+    <img src="../images/colorbar.gif" align="middle" width="540" height="4">
+  </center>
+  <p><i>Updated 5-18-99. &COPY; L. B. Cebik, W4RNL. Data may be used for personal purposes, but may not be reproduced for publication in print or any other medium without permission of the author.</i></p>
+  <center>
+    <img src="../images/colorbar.gif" align="middle" width="540" height="4">
+  </center><a href="../radio.html"><img src="../images/return.gif" align="bottom" width="40" height="40"> Go to Main Index</a><br>
+  <p class="topics-footer">Topics: <a href='../topics.html#dipole'>Dipole</a>, <a href='../topics.html#doublet'>Doublet</a>, <a href='../topics.html#hf-upper'>HF Upper</a></p><br>
+</body>
+</html>

content/wire/4.html

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+<!DOCTYPE html>
+<html lang="en">
+<head>
+  <meta charset="UTF-8">
+  <meta name="keywords" content="HF Upper, X Beam, Hex Beam, Moxon">
+  <title>A Tale of 4 Beams: The X, the Hex, the Square, and the Rect</title>
+  <link rel="stylesheet" href="../styles.css">
+</head>
+<body>
+  <center>
+    <h1>A Tale of 4 Beams: The X, the Hex, the Square, and the Rect</h1>
+  </center>
+  <center>
+    <h3>L. B. Cebik, W4RNL</h3>
+  </center><br>
+  <center>
+    <img src="../images/colorbar.gif" align="middle" width="540" height="4">
+  </center><br>
+  <br>
+  <p>From time to time, interest reemerges in some long-standing designs for compact planar (2- dimensional) beams. Unfortunately the interest seems to focus on a single design at a time rather than on the design as a member of a family of designs. Equally unfortunately, the interest usually stems from the publication of some peak performance figures for a particular design rather than from the antenna's performance across an entire band. Consequently, misunderstandings of antenna potentials multiply endlessly.</p>
+  <p>One of the family of beams whose members rouse periodic interest is the end-coupled clan. If the ends were connected, these would all make versions of a loop. However, with the ends spaced properly, each member forms a directional beam. Another apt name for the group might refer to the semi-closed geometry of the antennas. With closed loops, these antennas share the feature of tending toward larger dimensions with significant increases of element diameter.</p>
+  <p>Under any name, the family has two branches: those whose center structures form Vees that point at each other bottom to bottom and those whose centers parallel each other. Among the features that clan members have in common is a flat structure with an area that is just over 0.6 square wavelengths--in other words, about 1/4 by 1/4 wavelength. Hence, the lure of the family is its compact size.</p>
+  <p>It may be useful to explore the main members of the family individually to seek out their potential. I have selected 20 meters as the test band. To keep comparisons fair, I have constructed all models of #12 copper wire. However, some of the family members lend themselves to self-supporting aluminum tubing construction, and I shall note the potential performance changes that may result from building a tubing version of the antenna. The use of tubing for part or all of the structure, of course, will alter the dimensions from the ones used with the #12 wire versions.</p>
+  <p>The antennas that we shall examine are these:</p>
+  <ul>
+    <li>1. The folded X-beam</li>
+    <li>2. The hex beam</li>
+    <li>3. The VK2ABQ square</li>
+    <li>4. The Moxon rectangle</li>
+  </ul>
+  <p>As yet, I do not have any parallelograms, pentagons, or octagons in my collection.</p>
+  <p><b>The Folded X-Beam</b></p>
+  <center>
+    <img src="4-a.gif" width="389" height="393" border="2" vspace="5" hspace="9">
+  </center>
+  <p><b>Fig. A</b> shows the outlines of a folded X-beam. If you are interested in the history and details of the folded X-beam, see "Modeling and Understanding Small Beams: <a href="../mu/mu1a.html">Part 1: The X-Beam</a>," <i>Communications Quarterly</i>, 5 (Winter, 1995), 33-50. Ordinarily, the Vee portions of the folded X-beam are constructed of tubing supported by a center hub. Then wire tails for the driver and director are run from one corner toward the other, often taped to a perimeter cord that also holds the four arms in a fixed arrangement.</p>
+  <p>Modeling the usual construction of an X-beam is not feasible with NEC, since the program has an invariant tendency to yield inaccurate results with angular junctions of wires having different diameters. So, I have fashioned a model using #12 copper wire throughout. The performance differences are these: the all-wire version has a slightly lower maximum gain (by about 0.2 dB) and a slightly narrower 2:1 SWR bandwidth (about 50 kHz narrower) than the hybrid tubing/wire version. Incidentally, the hybrid version can be directly modeled with public-domain MININEC if one uses length-tapering toward the sharp angle corners.</p>
+  <p>Folded X-beams are normally designed for driver-director arrangements, since it is difficult to obtain significant performance with a driver-reflector arrangement. In the folded configuration of <b>Fig. A</b>, the parasitic element almost "wants" to be a director. In less metaphorical terms, a modestly performing driver-reflector design, with only a slight change of reflector length, will reverse its pattern and hold that reversal, even though the parasitic element is considerably longer than one might expect for a director. It is also possible to tune the director to move the peak front-to-back portion of the operating curve across the band. By lengthening the director and adding a remotely adjusted bit of capacitive reactance at the center, the peak performance region can be moved across an amateur band. However, the model used here employs a fixed construction, as the following table shows.</p>
+  <pre>X-Beam                                    Frequency = 14.1  MHz.
+
+Wire Loss: Aluminum -- Resistivity = 4E-08 ohm-m, Rel. Perm. = 1
+
+              --------------- WIRES ---------------
+Wire Conn. --- End 1 (x,y,z : in)  Conn. --- End 2 (x,y,z : in)   Dia(in) Segs
+1          -99.000, 17.000,  0.000  W2E1 -99.000, 99.000,  0.000    # 12   15
+2     W1E2 -99.000, 99.000,  0.000  W3E1  -6.000,  3.000,  0.000    # 12   25
+3     W2E2  -6.000,  3.000,  0.000  W4E1   6.000,  3.000,  0.000    # 12    3
+4     W3E2   6.000,  3.000,  0.000  W5E1  99.000, 99.000,  0.000    # 12   25
+5     W4E2  99.000, 99.000,  0.000        99.000, 17.000,  0.000    # 12   15
+6          -99.000,-11.000,  0.000  W7E1 -99.000,-99.000,  0.000    # 12   15
+7     W6E2 -99.000,-99.000,  0.000  W8E1  -6.000, -3.000,  0.000    # 12   25
+8     W7E2  -6.000, -3.000,  0.000  W9E1   6.000, -3.000,  0.000    # 12    3
+9     W8E2   6.000, -3.000,  0.000 W10E1  99.000,-99.000,  0.000    # 12   25
+10    W9E2  99.000,-99.000,  0.000        99.000,-11.000,  0.000    # 12   15
+
+              -------------- SOURCES --------------
+Source    Wire      Wire #/Pct From End 1    Ampl.(V, A)  Phase(Deg.)  Type
+          Seg.     Actual      (Specified)
+1           2     8 / 50.00   (  8 / 50.00)      1.000       0.000       V</pre>
+  <center>
+    <img src="4-1.gif" width="647" height="526" border="2" vspace="5" hspace="9">
+  </center>
+  <p>In <b>Fig. 1</b>, we find both the gain and front-to-back curves of the folded X-beam. Because the direction of the beam reverses between 14.3 and 14.35 MHz, the curves are cut off at 14.3 MHz. (The reversal to a driver-reflector beam yields only poor results, never reaching a 10 dB front-to-back ratio.) One of the inherent difficulties of the folded X-beam is that the maximum gain and the maximum front-to-back ratio are always separated in frequency. The gain at the maximum front-to-back peak is about 0.5 dB below peak. Both the gain and the front-to-back curves are quite steep, indicating a narrow operating passband, whatever the feedpoint impedance characteristics might be. In the past, the chief use of the folded X-beam has been on 10 meters as a home-brew project for those interested in the 28.3 to 28.5 MHz region of the band.</p>
+  <center>
+    <img src="4-2.gif" width="647" height="503" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The SWR curve, in <b>Fig. 2</b>, is referenced to 20 Ohms, which is approximately the impedance at the maximum front-to-back peak. Indeed, this design shows operating characteristics that are directly tied to the feedpoint impedance. A near-50-Ohm impedance is possible at the lowest frequency in the passband, with a low gain and relatively poor front-to-back ratio. Where the front-to-back ratio peaks, the impedance is from 20 to 25 Ohms, depending on the thickness of the element materials. At the maximum gain point, the feedpoint impedance drops to the 10-15-Ohm region. Wire versions of the antenna tend to show impedance values at the low end of the ranges indicated, while tubular and hybrid versions yield impedances values at the higher ends of the ranges.</p>
+  <center>
+    <img src="4-3.gif" width="640" height="380" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The peak gain and 180-degree front-to-back ratio figures can give a misimpression. The peak gain of about 6 dBi (free space) rivals that of a 2-element Yagi whose elements take twice the space side-to-side. Likewise, the peak 180-degree front-to-back ration of over 32 dB sounds impressive. However, the patterns in <b>Fig. 3</b> tell a somewhat different tale (as do the passband graphs we have viewed. An averaged front-to-rear ratio for the entire rear area of the beam has, within the 200 kHz of prime operation, a value of between 10 and 15 dB--no better than a common 2-element driver-reflector Yagi. The Yagi would also have superior gain over X-beam at every frequency and be able to cover the entire 20-meter band. A 2-element Yagi with about 1/8 wavelength element spacing and loaded elements that are about 3/4ths full size would occupy about the same area as the X-beam with broader performance curves. Hence, the folded X-beam has fallen into relative disuse.</p>
+  <p><b>The Hex Beam</b></p>
+  <center>
+    <img src="4-b.gif" width="385" height="397" border="2" vspace="5" hspace="9">
+  </center>
+  <p>If we fold the X-beam tails outward, we obtain the basic configuration of the hex(agon) beam, although true hex beams are built as closely to the hexagon geometry as the support structure will permit. <b>Fig. B</b> shows the outline of the model used to generate performance curves. The details of the model used in this study, which is a significantly modified version of a model originally provided by N7CL, follow in the chart.</p>
+  <pre>hex beam:  20 meters               Frequency = 14.1  MHz.
+
+Wire Loss: Aluminum -- Resistivity = 4E-08 ohm-m, Rel. Perm. = 1
+
+              --------------- WIRES ---------------
+Wire Conn. --- End 1 (x,y,z : in)  Conn. --- End 2 (x,y,z : in)   Dia(in) Segs
+1          -108.00, 19.500,  0.000  W2E1 -61.800,113.000,  0.000    # 12   22
+2     W1E2 -61.800,113.000,  0.000  W3E1  -9.950, 25.900,  0.000    # 12   26
+3     W2E2  -9.950, 25.900,  0.000  W4E1   9.900, 25.900,  0.000    # 12    5
+4     W3E2   9.900, 25.900,  0.000  W5E1  61.800,113.000,  0.000    # 12   26
+5     W4E2  61.800,113.000,  0.000       108.000, 19.500,  0.000    # 12   22
+6          -112.00,-12.900,  0.000  W7E1 -61.800,-113.00,  0.000    # 12   23
+7     W6E2 -61.800,-113.00,  0.000  W8E1  -9.950,-25.900,  0.000    # 12   26
+8     W7E2  -9.950,-25.900,  0.000  W9E1   9.950,-25.900,  0.000    # 12    5
+9     W8E2   9.950,-25.900,  0.000 W10E1  61.800,-113.00,  0.000    # 12   26
+10    W9E2  61.800,-113.00,  0.000       112.000,-12.900,  0.000    # 12   23
+
+              -------------- SOURCES --------------
+Source    Wire      Wire #/Pct From End 1    Ampl.(V, A)  Phase(Deg.)  Type
+          Seg.     Actual      (Specified)
+1           3     3 / 50.00   (  3 / 50.00)      1.000       0.000       I
+
+                -------- TRANSMISSION LINES ---------
+Line  Wire #/% From End 1   Wire #/% From End 1    Length       Z0   Vel Rev/
+      Actual  (Specified)   Actual  (Specified)                Ohms Fact Norm
+1      3/50.0  (  3/50.0)  Short ckt (Short ck)   12.000 in   600.0  1.00</pre>
+  <p>One feature of this model is the relatively wide spacing of the centers of the Vee-ed sections. This move tends to lower the feedpoint impedance to the 25-Ohm region, and the model uses a 12" stub of 600-Ohm shorted transmission line as a beta hairpin to effect a 50-Ohm match. It is possible to bring the center points of the driver and reflector closer together to obtain a direct 50-Ohm match. However, two deficits emerge with this move. First, the 50-Ohm match does not extend across the entire 20-meter band because the sharpness of the geometry yields a corresponding tuning sharpness. In contrast, the beta-matched 25-Ohm impedance does cover the entire 20-meter band with a 50-Ohm SWR of under 2:1. Second, with the center Vee points brought closer together, array performance smooths out across the band, but at much lower levels of gain and front-to-back ratio than we obtain from the wider-spaced center region. Therefore, I have chosen to look at the lower impedance version of the antenna with its better performance peaks.</p>
+  <p>The hex beam has a design affinity with a number of other members of the end-coupled clan that we shall not examine here. The slope of the outer sections of each end toward the other element is a property shared by several interesting antenna designs, including a 2-element reversible wire beam for 40 meters developed by AA2NN. The antenna uses a double slope, since the elements each form an inverted Vee. As well, each element end approaches the corresponding end of the other element. The result is a beam that requires only two center supports. As well, by using rope on the ends of the elements, the tie down points will also be reduced to 2. Equally related to the outer structure of the hex beam is the 3-element 40-meter reversible Yagi developed by WA3FET. It uses a linear driver and a pair of parasitic elements, each of which is sloped toward the end of the driver. One parasitic element is loaded for reflector duty. One advantage of element tips that slope toward each other rather than point directly at each other, is the greater ease of adjustment. Small changes of spacing of the tips produce less radical effects than when the tip are end-to-end.</p>
+  <center>
+    <img src="4-4.gif" width="648" height="526" border="2" vspace="5" hspace="9">
+  </center>
+  <p><b>Fig. 4</b> presents the gain and 180-degree front-to-back ratio figures across 20 meters. The gain variation across the band is nearly 2 dB, a fairly high figure among common 2-element beam designs. The front-to-back ratio shows a very sharp peak, but decreases rapidly to band-edge values in the 8 to 12 dB range. Peak operation of this antenna has a bandwidth of 100 to 150 kHz, with the remainder of the band showing relatively mediocre performance. Nonetheless, like all members of the semi-closed geometry family, the hex beam permits a high front-to-back peak whose decline is steeper below the peak frequency than above it.</p>
+  <center>
+    <img src="4-5.gif" width="648" height="497" border="2" vspace="5" hspace="9">
+  </center>
+  <p><b>Fig. 5</b> illustrates one of the illusions of SWR. One could suggest that this model of the hex beam antenna has an operating bandwidth that covers the entire band, since the 50-Ohm SWR is less than 2:1 across 20 meters. However, operating bandwidth involves more parameters than just the SWR. Evaluating the gain and front-to-back ratio is equally as important, if not more so, than the SWR. For this particular design of the hex beam, the only wide-band parameter is SWR. Gain and front-to-back ratio values are relatively narrow band properties.</p>
+  <center>
+    <img src="4-6.gif" width="640" height="380" border="2" vspace="5" hspace="9">
+  </center>
+  <p><b>Fig. 6</b> shows free-space azimuth patterns for the first 200 kHz of 20 meters. The pattern at 14.1 MHz is well controlled, but off peak, the rearward pattern spreads to average values in the 15 dB range. Beyond 14.2 MHz, the rearward pattern spreads larger and the forward gain decreases rapidly.</p>
+  <p>In general, like the X-beam and other beams based upon vee-ing the center parts of the elements, the hex beam shows a quite narrow operating bandwidth relative to gain and front-to-back ratio. The rate and total gain change across the band and the band-edge front-to-back ratio values are very important in evaluating the operating bandwidth of an antenna.</p>
+  <p>For further extensive information on home-brew hexbeams, see G3TXQ's website <a href="http://www.karinya.net/g3txq/hexbeam/" target="_blank">www.karinya.net/g3txq/hexbeam/</a> or K4KIO's site <a href="https://www.k4kio.com/" target="_blank">www.k4kio.com</a></p>
+  <p><b>The VK2ABQ Square</b></p>
+  <center>
+    <img src="4-c.gif" width="380" height="350" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The VK2ABQ Square (and the Moxon Rectangle) are more fully described in "Modeling and Understanding Small Beams: <a href="../mu/mu2a.html">Part 2: VK2ABQ Squares and The Modified Moxon Rectangle</a>," <i>Communications Quarterly</i>, (Spring, 1995), 55-70. The origins of the square go back to the 1930s, only to disappear and re-emerge in the 1960s. <b>Fig. C</b> shows the outlines of a modified square. The modification consists of loading the reflector with a shorted transmission line stub about 6" long to move the peak performance point without disturbing the square shape.</p>
+  <p>The original VK2ABQ square used very close-spaced element tips--only a literal coat button apart. However, very close tip spacing creates an array with narow-band properties, and small variations in construction can yield large variations in performance. Therefore, the model below uses fairly wide spacing (34") for the element tips.</p>
+  <pre>VK2ABQ 20 Meters                           Frequency = 14.15  MHz.
+
+Wire Loss: Copper -- Resistivity = 1.74E-08 ohm-m, Rel. Perm. = 1
+
+              --------------- WIRES ---------------
+Wire Conn. --- End 1 (x,y,z : in)  Conn. --- End 2 (x,y,z : in)   Dia(in) Segs
+1          -118.22, 16.889,  0.000  W2E1 -118.22,106.159,  0.000    # 12    6
+2     W1E2 -118.22,106.159,  0.000  W3E1 118.222,106.159,  0.000    # 12   13
+3     W2E2 118.222,106.159,  0.000       118.222, 16.889,  0.000    # 12    6
+4          -118.22,-16.889,  0.000  W5E1 -118.22,-106.16,  0.000    # 12    6
+5     W4E2 -118.22,-106.16,  0.000  W6E1 118.222,-106.16,  0.000    # 12   13
+6     W5E2 118.222,-106.16,  0.000       118.222,-16.889,  0.000    # 12    6
+
+              -------------- SOURCES --------------
+Source    Wire      Wire #/Pct From End 1    Ampl.(V, A)  Phase(Deg.)  Type
+          Seg.     Actual      (Specified)
+1           7     2 / 50.00   (  2 / 50.00)      1.000       0.000       I
+
+                -------- TRANSMISSION LINES ---------
+Line  Wire #/% From End 1   Wire #/% From End 1    Length       Z0   Vel Rev/
+      Actual  (Specified)   Actual  (Specified)                Ohms Fact Norm
+1      5/50.0  (  5/50.0)  Short ckt (Short ck)    5.892 in   600.0  1.00</pre>
+  <p>As the model shows, this version of the antenna is off square by about 12 inches. In this highly square (if imperfectly square) configuration, the feedpoint impedance is about 100 Ohms, making the antenna a candidate for a 2:1 balun at the feedpoint.</p>
+  <center>
+    <img src="4-7.gif" width="647" height="528" border="2" vspace="5" hspace="9">
+  </center>
+  <p>As shown by <b>Fig. 7</b>, the VK2ABQ square is a relatively low gain beam, although the gain varies only about 1.1 dB across the band. Hence, the 4.05 dB gain at the high end of the band equals that of the hex beam. The square's 180-degree front-to-back ratio peaks above 34 dB. Although the curves are fairly steep, the band edge values are about 15 dB--not bad for a 2-element parasitic beam that is about 1/4 wavelength on a side.</p>
+  <center>
+    <img src="4-8.gif" width="647" height="509" border="2" vspace="5" hspace="9">
+  </center>
+  <p>As <b>Fig. 8</b> shows, the real surprise of the modified VK2ABQ square is the 100-Ohm impedance curve. Across all of 20 meters, the resistive portion of the feedpoint impedance varies by under 6 Ohms, and the reactance varies by a similar amount. Hence, the SWR curve is very flat indeed. A 2:1 balun would permit operation across the entire 20-meter band with an exceptionally low SWR and no conditions to incur losses within the balun.</p>
+  <center>
+    <img src="4-9.gif" width="640" height="345" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The VK2ABQ was the basis for the later Moxon Rectangle. The key performance feature absorbed from the square was the excellent control of the rear portion of the radiation pattern. <b>Fig. 9</b> shows the band-edge and mid-band pattern for the square. If the square is constructed of 1" aluminum tubing, the band-edge front-to-back ratio improves to nearly 20 dB, with a small increase in array gain as well.</p>
+  <p>In all, the square is a relatively wide-band array whose characteristic remain reasonably level across the band (gain and impedance) or hold to minimal acceptable levels (front-to- back ratio). However, the chief deficit of the square is gain. In fact, one can preserve the front-to-back performance while improving gain--and as a bonus achieve a direct 50-Ohm match. The cost is going considerably out of square.</p>
+  <p><b>The Moxon Rectangle</b></p>
+  <center>
+    <img src="4-d.gif" width="407" height="249" border="2" vspace="5" hspace="9">
+  </center>
+  <p>Because the 3 family members we have so far examined use relatively wide spacing between facing element tips, many designers have ignored the effects of this dimension. The result has been a number of fairly poor designs. The element tip spacing influences the relative proportions of every other dimension of any of the family members. Nowhere is this more apparent than with the optimized Moxon rectangle, sketched in Fig. D. The combination of close tip coupling as well as more extended parallel element coupling allows the Moxon rectangle to recover the gain lost by the square while maintaining fairly wide-band operating characteristics. It is the longer sections of parallel elements that permit the close tip spacing to be controllable without sudden shifts in the direction of the pattern.</p>
+  <p>The #12 copper wire model for this study reveals that the side-to-side length is about 3/8 wavelength, while the front-to-back size is about 1/8 wavelength. Hence, the total area of the antenna is less than the 1/4 wavelength squares, although the turn radius is greater. The details of the model used here are as follows:</p>
+  <pre>Moxon rectangle                             Frequency = 14.175  MHz.
+
+Wire Loss: Copper -- Resistivity = 1.74E-08 ohm-m, Rel. Perm. = 1
+
+              --------------- WIRES ---------------
+Wire Conn. --- End 1 (x,y,z : in)  Conn. --- End 2 (x,y,z : in)   Dia(in) Segs
+1          -151.74, 64.188,  0.000  W2E1 -151.74,110.377,  0.000 8.08E-02   5
+2     W1E2 -151.74,110.377,  0.000  W3E1 151.740,110.377,  0.000 8.08E-02  35
+3     W2E2 151.740,110.377,  0.000       151.740, 64.188,  0.000 8.08E-02   5
+4          -151.74, 56.433,  0.000  W5E1 -151.74,  0.000,  0.000 8.08E-02   7
+5     W4E2 -151.74,  0.000,  0.000  W6E1 151.740,  0.000,  0.000 8.08E-02  35
+6     W5E2 151.740,  0.000,  0.000       151.740, 56.433,  0.000 8.08E-02   7
+
+              -------------- SOURCES --------------
+Source    Wire      Wire #/Pct From End 1    Ampl.(V, A)  Phase(Deg.)  Type
+          Seg.     Actual      (Specified)
+1          17     2 / 47.14   (  2 / 47.14)      0.707       0.000       V</pre>
+  <p>As the model shows, the rectangle is about 50% longer (side-to-side) than the squares. Tip-to-tip spacing is about 8". In the August, 2000, issue of <i>AntenneX</i>, I published a small program that inputs only the design frequency and wire diameter to yield optimized dimensions for Moxon rectangles for the HF and VHF regions. The designs provide a direct 50-Ohm match, whether used for rotatable or reversible beams.</p>
+  <center>
+    <img src="4-10.gif" width="645" height="527" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The gain curve in <b>Fig. 10</b> for the Moxon is a full dB better than for the square, although the total change in gain across the band is about the same. Since the Moxon rectangle can easily be fabricated of aluminum tubing, the result will be another 0.2 dB of gain and slightly less change in the gain across the band. As well, the band-edge front-to-back ratio values will improve to nearly 20 dB from the wire values of 15 dB. As with all of the semi-closed geometry designs, the front-to-back ratio is peaked just below the center of the band in order to achieve relatively similar front-to-back values at the band edges.</p>
+  <p>Both the square and the Moxon use the combination of parallel element coupling and end-coupling to achieve a very high front-to-back ratio at a design frequency. Indeed, in both cases, the current magnitude and phasing on the parasitic element center is very close to the precise values needed for a maximum front-to-back ratio if each element were to be independently fed and phased. Only the existence of the "tails," which radiate (if only weakly), prevents the pattern from becoming the deep dimple of a perfectly phased pair of elements.</p>
+  <center>
+    <img src="4-11.gif" width="646" height="508" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The 50-Ohm SWR curve in <b>Fig. 11</b> is for a direct match to coaxial cable with no matching required (although a common-mode current suppression choke or 1:1 balun is always in order). Unlike the SWR curve for the VK2ABQ square, the Moxon SWR curve shows a definite slope, although the band edge figures are acceptable under most conditions. The curve flattens further if one uses aluminum tubing of about 1" diameter for the antenna.</p>
+  <center>
+    <img src="4-12.gif" width="640" height="345" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The Moxon rectangle shares with the VK2ABQ square a nearly cardioidal pattern. The deepest "side" nulls do not occur at 90 degrees off the bearing of maximum gain, but somewhat further toward the rear, as is evident in <b>Fig. 12</b>. The rear lobes are well behaved, that is, they have no large quartering side lobes. The rearward lobes for the band edges shrink as the element diameter becomes larger.</p>
+  <p><b>Some Tentative Conclusions</b></p>
+  <p>This survey of semi-closed geometry end-coupled beams should suffice to reveal the family resemblances among the members of the clan. It may be useful to summarize some of the properties that both link and separate the individual members.</p>
+  <p>1. Designs with element center regions that are parallel or only gently sloped outward toward the ends tend to show wider-band characteristics than those whose element centers are Vee-ed toward each other.</p>
+  <p>2. Element tips display two regions of coupling. Wider spacing between tips tends to produce lower gain, although small changes in spacing yield less radical effects. Closely spaced tips tend to be more critical and may be effectively usable only if most of the element length is either parallel or only gently slopes to bring the tips closer together.</p>
+  <p>3. Semi-closed beam designs tend toward loop properties, such as an increase in perimeter dimensions with an increase in element diameter. Sloping element designs are most immune to this effect and may show more typical linear element properties.</p>
+  <p>4. Designs that strive for a minimum turning radius tend either to have narrow-band characteristics or lower gain. The Moxon rectangle represents a compromise geometry that achieves as good or better gain than the other 2-element members of the clan while achieving a high front-to-back ratio and relatively broad-band characteristics. Sometimes the best square is a rectangle.</p>
+  <p>5. Both the front-to-back ratio and SWR curves tend to deteriorate much faster below the design frequency than above it. Therefore, to achieve relatively equal performance at both the lower and upper band edges, the appropriate design frequency is about 1/3 the way up the band. For 2-element driver-reflector designs, whether using a standard Yagi configuration or one of the end-coupled designs, the gain will decrease as frequency increases.</p>
+  <p>I have over the years built and used most of the designs we have discussed here in 10- meter versions, using both wire and aluminum construction. The models employed here are variants of those antennas, as well as of published data. No commercial antennas are modeled for these notes. Their intent is simply to show both the resemblances and differences among members of the end-coupled clan of beams.</p>
+  <center>
+    <img src="../images/colorbar.gif" align="middle" width="540" height="4">
+  </center>
+  <p><i>Updated 9-1-2000. &COPY; L. B. Cebik, W4RNL. The original item appeared in</i> AntenneX <i>for August, 2000. Data may be used for personal purposes, but may not be reproduced for publication in print or any other medium without permission of the author.</i></p>
+  <center>
+    <img src="../images/colorbar.gif" align="middle" width="540" height="4">
+  </center><a href="../radio.html"><img src="../images/return.gif" align="bottom" width="40" height="40"> Go to Main Index</a><br>
+  <p class="topics-footer">Topics: <a href='../topics.html#hf-upper'>HF Upper</a>, <a href='../topics.html#hex-beam'>Hex Beam</a>, <a href='../topics.html#moxon'>Moxon</a>, <a href='../topics.html#x-beam'>X Beam</a></p><br>
+</body>
+</html>

content/wire/40h.html

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+<!DOCTYPE html>
+<html lang="en">
+<head>
+  <meta charset="UTF-8">
+  <meta name="keywords" content="HF Lower, Dipole, Yagi, Loading">
+  <title>Half-Length Dipoles (for 40 Meters)</title>
+  <link rel="stylesheet" href="../styles.css">
+</head>
+<body>
+  <center>
+    <h1><img src="../images/wirecad.gif" align="middle" width="125" height="62" border="2" vspace="5" hspace="9">Half-Length Dipoles (for 40 Meters)<img src="../images/w4rnlwwr.gif" align="middle" width="99" height="83" border="2" vspace="5" hspace="10"></h1>
+  </center>
+  <center>
+    <h3>L. B. Cebik, W4RNL</h3>
+  </center>
+  <center>
+    <img src="../images/colorbar.gif" align="middle" width="540" height="4">
+  </center>
+  <p>Below 20 meters, full-length dipoles (and other antennas based on the dipole) present space problems. For many amateurs, such antennas are simply too long to fit within the modern urban and suburban yards. So the antenna builder begins to think of ways to shorten the dipole. The questions surrounding shortened antennas are complex. Some involve performance levels compared to the full-length dipole. Others concern the relative efficiency of antennas that use different means of shortening. Another group of questions focus on the mechanical issues created by various methods of shortening. Moreover, there are auxiliary matters, such as matching the shortened antenna to one of the standard feedlines in common use.</p>
+  <p>To explore these questions in a somewhat systematic manner, we shall pick a single antenna length on a single amateur band. 40 meters (7.0 to 7.3 MHz in the U.S.) is handy, since the average dipole length is in the vicinity of 67', just on the verge of fitting or not fitting a typical back yard. Let's use a half-length dipole and set its length at a fixed value of 33' for our explorations. Most of our antennas will use AWG #12 (0.0808" diameter) copper wire. With these simple premises, we can examine a myriad of ways of shortening dipoles, including but not limited to, folding back the elements, using inductive loads at the dipole center or along the element length, using end "hat" loads or element extensions, and employing U and other shapes. Each alternative method of shortening the length of a dipole has its own cluster of variations, its own set of issues, and its own set of consequences, for example, when constructing basic parasitic beams with half-length elements</p>
+  <p>This series of PDF documents tries to explore the basic territory in an orderly fashion, beginning with the development of basic properties of full size elements to form a baseline with which we can compare various techniques of shortening. The next step is to look at the performance potential of shortened elements and explore a few ways in which amateurs have simply folded and bent full size elements to fit a half-size space. Actual shortening takes us into the realm of element loading using inductive and other methods to resonate a half-length element. The last step is to look at some consequences of element shortening for the design and performance of directional beams.</p>
+  <p>Since each new step makes use of information developed in the previous step, you may wish to keep earlier episodes open while reading the new one. To return to this master page, use your browser's "previous page" button, since PDF documents have no ready means for returning to this page. The titles for each of the episodes should by self-explanatory.</p>
+  <p><a href="40ha.pdf">Part 1: The Full-Length Standard</a></p>
+  <p><a href="40hb.pdf">Part 2: Shortening and Reshaping the Dipole</a></p>
+  <p><a href="40hc.pdf">Part 3: Element Loading to Achieve Dipole Resonance</a></p>
+  <p><a href="40hd.pdf">Part 4: Basic Loaded-Element Parasitic Beams</a></p>
+  <p>Most of the material has appeared in other articles at this site, but under a scattered collection of specific topics and at different test frequencies. Drawing the ideas together may provide you with a more cohesive understanding of the principles involved in radically shortening a dipole and obtaining the best possible performance from it.</p>
+  <center>
+    <img src="../images/colorbar.gif" width="591" height="3">
+  </center>
+  <p><i>Updated 12-24-2007. &COPY; L. B. Cebik, W4RNL. Data may be used for personal purposes, but may not be reproduced for publication in print or any other medium without permission of the author.</i></p>
+  <center>
+    <img src="../images/colorbar.gif" width="591" height="3">
+  </center>
+  <p><a href="../radio.html"><img src="../images/return.gif" align="bottom" width="40" height="40"> Return to Amateur Radio Page</a></p>
+  <p class="topics-footer">Topics: <a href='../topics.html#dipole'>Dipole</a>, <a href='../topics.html#hf-lower'>HF Lower</a>, <a href='../topics.html#loading'>Loading</a>, <a href='../topics.html#yagi'>Yagi</a></p><br>
+</body>
+</html>

content/wire/40star.html

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+<!DOCTYPE html>
+<html lang="en">
+<head>
+  <meta charset="UTF-8">
+  <meta name="keywords" content="HF Multi Band, Horizontal Loop">
+  <title>A 40-Meter Star</title>
+  <link rel="stylesheet" href="../styles.css">
+</head>
+<body>
+  <center>
+    <h1><img src="../images/wirecad.gif" align="middle" width="125" height="62" border="2" vspace="5" hspace="9">A 40-Meter Star<img src="../images/w4rnlwwr.gif" align="middle" width="99" height="83" border="2" vspace="5" hspace="10"></h1>
+  </center>
+  <center>
+    <h3>L. B. Cebik, W4RNL</h3>
+  </center>
+  <center>
+    <img src="../images/colorbar.gif" align="middle" width="540" height="4">
+  </center>
+  <p>The horizontally oriented 1 wavelength square loop is a fairly standard low-HF amateur antenna. It lends itself to use with parallel feedline for multi-band application. However, a 1 wavelength loop tends to radiate broadside to the loop. Therefore, the antenna tends to provide better performance on bands above the lowest.</p>
+  <h4>The Standard Square and the 4-Pointed Star Loops</h4>The need for a longer circumference is often at odds with amateurs who have only limited space for wire antennas on the lower HF bands. However, one way to increase the circumference of a loop without increasing its footprint is to draw in the 4 sides of the loop toward the center. The result is a 4-pointed star configuration. <b>Fig. 1</b> shows the difference between the standard and star loops, as viewed from above (or below, as the case may be).
+  <center>
+    <img src="40star-1.gif" width="520" height="350" border="2" vspace="5" hspace="9">
+  </center>
+  <p><b>Fig. 1</b> also provides us with a key to the main dimensions of the loop and the star. The length of a side for a square horizontally oriented loop is also the length of one side of its footprint. For the 40-meter (7.15 MHz) test case, each side of the loop is about 36.2' long for a near resonant loop. This provides an antenna and a footprint circumference of 144.8' or about 1.05 wavelengths at 7.15 MHz for a near-resonant loop. On the right side of <b>Fig. 1</b> is the star. Here, we must distinguish between the wire length and the footprint. For a near resonant loop, we require a footprint side dimension of about 31.9', which results in a footprint circumference of 127.6'. This dimension set is actually smaller than for the square loop. However, as shown in the sketch, each wire is stretched inward toward the center. We cannot make the wire touch at the center, but we can come in rather close. The most radically inset case that I have so far explored positions the apex of each angle formed from the side wires at 1.75' from the antenna center. This yields a distance of about 3.5' between opposing points. The resulting wire length for each side of each point in the star is about 21.35'. The total wire circumference thus becomes about 170.8' or close to 1.25 wavelengths.</p>
+  <p>We can compare the potential performance of the two configurations on 40 meters via the following table of modeled results.</p>
+  <pre> . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
+              Square and Star Loop Performance at 7.15 MHz
+
+Antenna Height 50'.  Antenna Wire AWG #12 copper.  "Insets" refers to the
+distance of the limit of the star side inset point from the exact center
+of the array.
+
+                      Gain       El. Angle        Feed Z
+                      dBi        Degrees          R +/-jX Ohms
+Square                5.54       47               157.5 - j 6.3
+Star: 1.75' insets    5.50       39                65.7 + j 9.0
+Star: 2.0' insets     5.50       39                66.8 + j12.0
+Star: 3.0' insets     5.50       40                71.1 - j 0.6
+ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .</pre>
+  <p>Several aspects of the tabular data are significant. First, the 40-meter gain of the two versions of the loop is virtually the same. However, the elevation angle of maximum radiation is considerably lower in the star version. <b>Fig. 2</b> graphically illustrates these matters by showing the two azimuth patterns, each at is respective TO angle, to exactly overlay each other. However, the elevation pattern of the star along the axis of maximum radiation has a noticeably lower angle of maximum radiation (take-off or TO angle).</p>
+  <center>
+    <img src="40star-2.gif" width="400" height="240" border="2" vspace="5" hspace="9">
+  </center>
+  <p>Second, if operation is contemplated only on 40 meters, then the impedance of the star configuration is suitable for a coaxial cable as the feedline Either 50-Ohm or 75-Ohm cable will do. For similar operation, the square configuration would require either the use of a parallel feedline or the use of a 4:1 balun with a 50-Ohm coaxial cable feedline.</p>
+  <p>Third, the star configuration is not especially sensitive to just how far toward the array center we push the insets. The distances from center shown may be doubled to see how far apart we may place the inner points of the star. There is considerable room for variation before we lose our advantage over the square loop in terms of TO angle. However, note that the 3.0' inset has bumped the TO angle upward one notch. As we further move the inner start points away from center, the antenna slowly returns to the characteristics of a simple square loop.</p>
+  <p>The principle behind the star is an attempt to increase its wire circumference length without increasing its footprint. The 0.2- wavelength increase, while not giving us the almost pure edge-wise radiation of a 2 wavelength loop, does raise the entire wire length in the star loop to 1.25 wavelengths. That much length is sufficient to lower the 40-meter radiation angle by a noticeable amount.</p>
+  <h4>The Square Loop as a Multi-Band Wire Antenna</h4>The square loop allows us to feed the antenna on higher bands, relative to the base-line 40-meter band to which we have cut it. We can summarize the performance with the following tabulated samples for the HF bands.
+  <pre> . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
+                    40-Meter Square Loop Performance
+
+Antenna Height:  50'.  Antenna Wire AWG #12 copper.
+
+Freq.      Gain       TO angle   Feed Z           Pattern Shape
+MHz        dBi        Degrees    R+/-jX Ohms
+7.15       5.5        47          160 - j    6    Oval
+10.125     4.8        32         3060 + j 3140*   Almost square
+14.1       8.5        19          275 + j  120    4-leaf clover
+18.1       7.2        16         1035 + j 1480*   wobbly oval
+21.1       8.7        13          255 + j   55    4 main lobes, 60
+                                                  degrees off axis
+24.95      8.0        11         1230 - j 1380*   6 near-equal lobes
+28.1       10.8       10          265 + j  115    4 lobes 45 degrees
+                                                  off axis
+ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .</pre>
+  <p>With exceptions, the patterns generally are strongest in a line through the feedpoint and the corresponding center point of the wire opposite. We may call this the main axis of the antenna. On two bands of high interest, however, the patterns depart from the noted tendency.</p>
+  <center>
+    <img src="40star-3.gif" width="400" height="240" border="2" vspace="5" hspace="9">
+  </center>
+  <p><b>Fig. 3</b> shows the azimuth patterns of the square loop on 15 and 10 meters, with the axis presumed to run vertically on the page. The 15-meter pattern forms a sort of butterfly, with small lobes along the antenna axis. However, the strongest lobes are angled to the sides by about 60 degrees. The 10-meter pattern has only 4 notable lobes, each about 45 degrees off axis.</p>
+  <p>We may also note in passing the starred entries in the feedpoint impedance (Feed Z) column. Each of the non-harmonic bands presents an impedance where the resistance and the reactive components are both above 1000 Ohms. Without careful attention to the characteristic impedance and length of the parallel feedline used, the impedance at the antenna tuner terminals may fall outside the range of values that it can match.</p>
+  <h4>The 4-Pointed Star Loop as a Multi-Band Antenna</h4>We may perform the same modeling experiment with the 4-pointed star loop to evaluate its potential as a multi-band antenna for 40-10 meters. The results appear in the following table.
+  <pre> . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
+                     40-Meter Star Loop Performance
+
+Antenna Height:  50'.  Antenna Wire AWG #12 copper.
+
+Freq.      Gain       TO angle   Feed Z           Pattern Shape
+MHz        dBi        Degrees    R+/-jX Ohms
+7.15       5.5        39           65 + j   10    Oval
+10.125     6.7        26         6820 - j 7650*   Diamond
+14.1       9.3        19          540 + j 1850*   4-leaf clover
+18.1       6.9        16          925 + j   75    Broad beam:  F-B 5.2 dB
+21.1       6.2        13          945 - j 1270*   Broad beam:  F-B 1.3 dB
+24.95      6.8        11           55 + j  340    Broad beam:  F-B 2.5 dB
+28.1       6.9        10          715 - j  670    Triple forward lobes
+ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .</pre>
+  <p>For the entries called "Broad beam," the direction of maximum gain is toward the side of the star containing the feedpoints. If we overlay the outline of the antenna on top of the azimuth patterns in <b>Fig. 4</b>, the feedpoint will be above the plot center line across the page.</p>
+  <center>
+    <img src="40star-4.gif" width="600" height="440" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The patterns show one potential advantage of the star as a multi-band antenna. On all bands, there is a main lobe along the antenna axis through the feedpoint. Hence, the user is always aware of the direction of strongest signal. (30 meters is the one exception, but the main lobe to the reverse of the feedpoint side is only 0.7 dB stronger than on the feedpoint side, a difference that will not be detectable in operation.) Although the beam action--that is, having a small front-to-back ratio--is small, the reliability of having the main lobe along the same axis on every band used is a distinct plus.</p>
+  <p>There are three bands on which the reactance rises above 1000 Ohms. However, only on 30 meters are the values for both resistance and reactance so high as to create a very distinct problem for matching the feedline termination to the transceiver 50-Ohm system.</p>
+  <h4>Why?</h4>The distinctness of the square loop and the star loop patterns should arouse some curiosity as to the reason for the differences. <b>Fig 5</b> provides a partial answer.
+  <center>
+    <img src="40star-5.gif" width="540" height="440" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The upper diagrams compare the relative current magnitude distribution of the two loops on 40 meters. The current on the star remains higher further outward toward the array corners than on the square loop, and this phenomenon plays a role in lowering the elevation angle of maximum radiation (the take-off or TO angle). Otherwise, the gain and pattern shape of the 2 versions of the loop are the same.</p>
+  <p>The 15-meter case is especially interesting. For the star loop, the current magnitude peaks and valleys appear in close proximity along the outward star-point wires. Hence, the currents (or, more properly, the fields that result) tend to simply add to or subtract from each other-- with due place given to the phase of each current magnitude sampled. However, in the square loop, we have current magnitude peaks more linearly separated from each other, with distinct peaks at the four corners of the array. The result is the 6-lobes pattern, with the largest lobes at a considerable angle from the axis of the antenna.</p>
+  <p>These brief notes suggest that for some users of square loops, modification to a star design may be useful. The array dimensions for 40 meters will easily scale to 80 and 160 meters, although most users will have difficulty in scaling the height as well as the wire length. Since we are only approximating resonance on the lowest band of use and presuming parallel feedline to an antenna tuner, fussiness with dimensions seems out of place. Since the wire of the antenna has a small diameter relative to a wave length, any 50-Ohm resonance on the lowest band of use is likely to be a very narrow-band phenomenon.</p>
+  <p>Nonetheless, for the loop-user who wishes a lower TO angle on the lowest band of use and a pattern that has a maximum along the axis of the antenna on every band used, the 4-point star is viable alternative to the standard square loop. The cost is less than 20% more wire, which is likely to be the cheapest part of the antenna anyway. The star loop is not an answer to every loop problem. However, it does show that it pays to explore different wire geometries to see whether they have any potential for use.</p>
+  <center>
+    <img src="../images/colorbar.gif" align="middle" width="540" height="4">
+  </center><br>
+  <br>
+  <p><i>Updated 04-13-2003. &COPY; L. B. Cebik, W4RNL. Data may be used for personal purposes, but may not be reproduced for publication in print or any other medium without permission of the author.</i></p>
+  <center>
+    <img src="../images/colorbar.gif" align="middle" width="540" height="4">
+  </center>
+  <p><a href="../radio.html"><img src="../images/return.gif" align="bottom" width="40" height="40"> Go to Main Index Page</a><br></p>
+  <p class="topics-footer">Topics: <a href='../topics.html#hf-multi-band'>HF Multi Band</a>, <a href='../topics.html#horizontal-loop'>Horizontal Loop</a></p><br>
+</body>
+</html>

content/wire/40va.html

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+<!DOCTYPE html>
+<html lang="en">
+<head>
+  <meta charset="UTF-8">
+  <meta name="keywords" content="HF Lower, Vertical">
+  <title>Screening 40-Meter Vertical Arrays</title>
+  <link rel="stylesheet" href="../styles.css">
+</head>
+<body>
+  <center>
+    <h1>Screening 40-Meter Vertical Arrays</h1>
+  </center>
+  <center>
+    <h3>L. B. Cebik, W4RNL</h3>
+  </center><br>
+  <center>
+    <img src="../images/colorbar.gif" align="middle" width="540" height="4">
+  </center>
+  <p>The lower HF amateur bands tend to feature vertical antennas. We find a few horizontal beams, especially on 40 and 30 meters, but they tend to be very large, heavy, and expensive. In contrast, vertical arrays make use of wire and usually use much simpler construction. Hence, their maintenance requirements are also simpler.</p>
+  <p>At 40 meters (and by extension and scaling 30 meters as well) the vertical dipole becomes feasible, especially if we find a way to shorten it somewhat without losing significant performance. From the vertical dipole, we may create a large number of array types. In these notes, I want to examine in order of increasing performance capabilities a collection of vertical dipole beams and arrays. So that all comparisons will be fair, every vertical array will use AWG #12 copper wire, perhaps the most common material for amateur wire antennas. As well, all antennas will be over average soil (conductivity 0.005 S/m, relative permittivity 13). The performance of vertical antennas will change with the quality of the soil beneath them and in the far-field reflection zone, but the changes will tend to be consistent within any given soil type. Therefore, if we know the performance of an array over average soil and the relative performance for a simple vertical dipole over both average soil and the specific soil type for a given installation, we may extrapolate the performance values for a more complex antenna system.</p>
+  <p><b>Vertical Dipole Basics</b></p>
+  <p>The root antenna for what follows might seem to be a vertical dipole. <b>Table 1</b> and <b>Fig. 1</b> summarize the properties of a vertical dipole that extends from 1' above ground level to a top height of 67.6'. The total wire length is 66.6', but this exact value applies only to a resonant vertical dipole over average soil. Slight adjustments are likely for different soil types, especially since the lower end of the antenna is so close to the ground. As will be the case for all models used in these comparisons, the performance and dimension values do not account for the influence of objects in the immediate area of the antenna installation, that is, the so-called ground clutter. All vertical antennas require as much clearance from ground clutter as the installation site will permit.</p>
+  <center>
+    <img src="40va-t1.gif" width="234" height="170" border="2" vspace="5" hspace="9">
+  </center>
+  <center>
+    <img src="40va-1.gif" width="300" height="320" border="2" vspace="5" hspace="9">
+  </center>
+  <p>For this and all following antennas, the test frequency will be 7.15 MHz. The clutterless radiation patterns show very normal characteristics--a circular azimuth pattern and a single elevation lobe with a low take-off (TO) angle. Hence, despite the low gain of the antenna, it finds general favor for its coverage and for its insensitivity to high-angle noise and signals. The simple outline sketch shows the current magnitude distribution along the antenna wire. One reason why the antenna patterns show a low TO angle stems from the relatively high position of the feedpoint or the region of highest current magnitude, just about 1/4 wavelength above ground. However, the close proximity to the ground at the lower end of the dipole does elevate its feedpoint impedance--from an expected 70-Ohm value up to the 98-Ohm value reported by NEC-4 for the sample antenna.</p>
+  <p>Many amateurs do not have 70' supports for a 40-meter vertical dipole in its simplest form. Individual circumstances vary, but let's suppose that the maximum support height is about 50'. Within this height restriction, we may still install a modified vertical dipole. Rather than accept the losses that center-loading or mid-element loading might create, we shall use end hats. An end hat or cap is a symmetrical structure at right angles to the main plane of the antenna. Radiation from the wires of the end cap largely self-cancels, leaving us with a vertical antenna in terms of the radiation patterns. Since the self-canceling portion of the radiation occurs in the low-current regions of the antenna, we preserve most of the performance that we might obtain from a full-length dipole. <b>Fog. 2</b> shows the outlines and current distribution on one version of such a T-capped dipole/</p>
+  <center>
+    <img src="40va-2.gif" width="360" height="320" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The end hats on a vertical dipole can use any symmetrical arrangement. The more radial arms that we create, the shorter each one must be for a given length of vertical wire in the center. However, many-spoked hats set up very significant support requirements. The T-cap requires only two wires at each end of the dipole, and we may run these wires along the non-conductive ropes that we often use to support wire vertical dipoles between two posts or trees. Reducing the number of hat wires to only two per end does not affect the performance relative to using greater numbers of hat wires. For example, the azimuth pattern of the T-cap dipole in the sketch shows only 0.03-dB of gain variation as we check all 360 degrees of the horizon.</p>
+  <p>The T-cap dipole that <b>Fig. 2</b> shows is only 35.5' long, stretched from 5' to 40.5' above ground. (One might easily raise the antenna by another 4 to 5 feet and still remain below the 50' ceiling that we set. The added height would also increase safety by raising the base wires with their high RF voltages above the level that family, friends, or even pets might touch.) The vertical section, then, is just over 1/4 wavelength. For all following 40-meter vertical antennas, we shall use the vertical section of the T-cap dipole. The two horizontal wires at the top and bottom of the antenna, various called arms or legs, are each exactly 10' long in the sample. Hence, the total width of the antenna is 20'. This width falls well within the clear area that we should have for any type of vertical dipole.</p>
+  <p>If we need to make adjustments for obtaining resonance in subsequent arrays that use the T-cap dipole, we shall adjust the length of the legs of the T, thereby leaving the center vertical section intact. The dimensions for the basic T-cap dipole appear in <b>Table 2</b>, along with some basic performance data. Within that data, only the first section is immediately relevant for comparison with the full-length dipole. First, note the 22-degree TO angle. This value is higher than the value for the full-length dipole, but the T-cap vertical's feedpoint is considerably lower (about 0.16 wavelength above ground). Partly as a function of the lower feedpoint height and partly as a function of the vertical-element shortening, the gain of our T-cap dipole is about 0.4-dB lower than the gain of the full-length dipole. Yet the T-cap dipole, as set up in the sample model, is nearly 30' lower in overall height, yielding what is for most amateur operators a much more manageable construction and maintenance situation.</p>
+  <center>
+    <img src="40va-t2.gif" width="329" height="303" border="2" vspace="5" hspace="9">
+  </center>
+  <p>In the table, we also find some performance values related to improvements that we may make in the local ground immediately beneath the vertical dipole. When we think of local ground improvements together with virtually any vertical antenna, most amateurs immediately think of radials. However, unlike the radials of a monopole, the radials beneath a vertical dipole (either full length or shortened) perform no antenna-completing function. Rather, they simply function to raise the conductivity of the soil immediately beneath the antenna. In cases 2 through 4 in the table, I created radial systems with the hub directly below the antenna. The 1/4 wavelength radials use AWG #12 wire buried 1' deep in the average ground. As the table shows, adding 4 radials amounts to wholly wasted effort, since the gain increase is only 0.04 dB. 16 radials provide a 0.3-dB gain increase, perhaps a marginal amount, considering the work involved. If radial installation is easy, we may increase the field to 64 radials and obtain nearly 0.7 dB gain increase. <b>Fig. 3</b> provides a view of the 3 radial field along with the T-cap dipole above them. You may estimate the installation work from the sketches.</p>
+  <center>
+    <img src="40va-3.gif" width="670" height="265" border="2" vspace="5" hspace="9">
+  </center>
+  <p>Note that none of the radial fields changes the TO angle of the elevation pattern. The TO angle is mostly a function of the far-field reflection zone, with is mostly well outside the radial limits. For the exercises, I did not change the antenna dimensions. Therefore, the major influence of the fields appears in the feedpoint impedance listings. As the size of the radial field increases, the resistive component of the impedance decreases and the reactance becomes more inductive. However, none of the changes in cases 2 through 4 present any operational concerns.</p>
+  <p>The last case presents an alternative method of local ground improvement. Instead of using a radial system, I modeled a wire-grid system below ground to simulate laying a screen of some sort below the antenna. The screen is square and 70' on a side. <b>Fig. 4</b> overlays the 16-radial system on the screen to show the change in ground coverage in the screen corner region. (I used the 16-radials so as not to obscure the grid. However, the most relevant comparison would be between the screen and the 64-radial system.)</p>
+  <center>
+    <img src="40va-4.gif" width="360" height="255" border="2" vspace="5" hspace="9">
+  </center>
+  <p>With the screen in place, we obtain almost 1.1 dB gain improvement over untreated soil, and about 0.4-dB improvement over the 64-radial system. The cost of the gain is a 1-degree increase in the TO angle and further drift in the feedpoint impedance. For antennas--like vertical dipoles--that do not require radials to function as the lower half of a dipole, screens may sometimes be the easiest and most effective means on improving soil quality immediately below the antenna. However, they do not provide that same benefits as living over very good soil, since the treatment does not also apply to the far-field reflection zone. For example, local area treatment may reduce ground losses below the antenna, but that results in a higher TO angle, because the improvement increases local reflection almost straight upward. The lower TO angles that we often associate with the same antenna over very good soil results from improved soil conductivity at considerable distances from the antenna.</p>
+  <p>In the end, soil improvement is a matter for the antenna user to decide after measuring the anticipated performance improvements against the amount and cost of work involved in local ground treatment. For the remainder of these notes, we shall use untreated soil beneath the antenna, so that the performance values in case 1 in <b>Table 2</b> become the reference points for all that follows.</p>
+  <p>Before we leave the basic T-cap vertical dipole, we should introduce one more set of considerations. We are interested not only in the performance of any antenna at the test frequency, but also across the entire band. 40 meters is a wider (but not the widest) amateur allocation, with a 4.2% bandwidth. Some antennas will handle the entire band; others will not. A basic dipole--either full-length or T-capped--shows only a show change of gain across the band. The case-1 T-cap dipole, for example, changes gain by only 0.07 dB from 7.0 to 7.3 MHz. Note that our performance concerns include not only the SWR properties, shown in <b>Fig. 5</b>, but also such matters of forward gain and front-to-back ratio, both of which will become important as we add directionality to the basic dipole.</p>
+  <center>
+    <img src="40va-5.gif" width="416" height="265" border="2" vspace="5" hspace="9">
+  </center>
+  <p>Although the resonant impedance of the basic T-cap dipole is in the vicinity of 75 Ohms, even the 50-Ohm SWR is below 2:1 from one band edge to the other. Hence, we may feed the dipole using either 50-Ohm or 70-Ohm coaxial cable. As usual, one should route the cable at right angles to the vertical dipole for as far as possible, providing supports so that the cable weight does not unduly stress the wire antenna. As well, I recommend the use of 2 common-mode current attenuators, one at the feedpoint to also serve as a balun, and the other at the entry to the operating building to attenuate any currents induced along the cable run.</p>
+  <p><b>A Bi-Directional Pair of T-Cap Dipoles</b></p>
+  <p>Because many of the arrays to follow will use a pair of T-cap dipoles, we should spend a moment two see what happens when we place a pair of these dipoles at a spacing of 1/2 wavelength (about 68.8') and feed them in phase. Because the in-phase fed pair of antennas will interact (or, otherwise put, exhibit mutual coupling), I extended the T arms to 10.31' each. As a result, the total array width, from outer T-tip to outer T-tip is about 89.4'. As shown in <b>Table 3</b> and the outline portion of <b>Fig. 6</b>, the array height above ground has not changed relative to using a single T-cap dipole.</p>
+  <center>
+    <img src="40va-t3.gif" width="574" height="182" border="2" vspace="5" hspace="9">
+  </center>
+  <center>
+    <img src="40va-6.gif" width="360" height="470" border="2" vspace="5" hspace="9">
+  </center>
+  <p>Perhaps the most surprising aspect of the array is the 4.1-dB gain improvement over a single T-cap vertical dipole. Of course, the added gain comes at a cost in the available beamwidth, now down to about 62 degrees. Nevertheless, the array is no taller and only a bit wider than a half-square, but provides more gain. The gain that is competitive with a bobtail curtain. In fact, the in-phase-fed pair of vertical dipoles lies at the theoretical core of all SCV (self-contained vertical) antennas, since the basic versions--ranging from deltas to rectangles to half-squares, all place two elements in phase, but at a spacing limited by the single-wire, single-feedpoint configuration. For further information on SCVs, see <a href="../scv/scv0.html">"Self-Contained Vertically Polarized Wire Antennas: A Family Album"</a></p>
+  <p>The independent feedpoint impedances of the two T-cap dipoles is about 56 Ohms. Therefore, we may equip the phased pair with a common feedpoint by using equal lengths of 70-Ohm feedline to a center point. Since the physical distance between each element's feedpoint and the center point is 1/4 wavelength, and since all 70-Ohm transmission lines have a velocity factor (VF) of well under 1.0, you may need to use 3/4 wavelength section to arrive at the required 100-Ohms at the junction, a value that becomes a good 50-Ohm match in a parallel connection. The lower portion of <b>Fig. 6</b> provides the modeled 50-Ohm SWR curve. The array covers the entire 40-meter band with an SWR value that is less than 2:1.</p>
+  <p>We shall not linger over the in-phase-fed pair of T-cap dipoles and their broadside bi-directional pattern. However, we shall occasion to mention them once more before we close our screening survey. Nevertheless, we should call to attention one more time the gain value produced by the pair of elements. The maximum gain in each direction will exceed the forward gain value that we may achieve from some of the more basic beams that we consider when thinking about a pair of T-cap dipoles and their best use.</p>
+  <p><b>Parasitic Driver-Reflector Beams Using T-Cap Dipoles</b></p>
+  <p>For many operating needs, a front-to-back ratio of at least 10 dB may be more important than additions to the arrays forward gain. In such cases, we may create an endfire array using standard 2-element Yagi principles. In a Yagi with full-length elements, the reflector is normally longer and the driver normally shorter than a freestanding resonant dipole. In creating the driver for a useful Yagi-type beam, we may simply reduce the T-legs to an individual length of 9.6' (for a total element width of 19.2').</p>
+  <p>However, we shall not stop to develop a reflector for the T-cap array that uses a larger element. Instead, we shall simply load the reflector and use the same dimensions as for the driver. Reflector loading of a 2-element parasitic array normally does not reduce the forward gain, and it may actually improve the front-to-back performance over a full size reflector. <b>Table 4</b> supplies the dimensions of the array, which spaces the elements 21' apart. <b>Fig. 7</b> shows the outline and the radiation patterns that we might expect.</p>
+  <center>
+    <img src="40va-t4.gif" width="611" height="179" border="2" vspace="5" hspace="9">
+  </center>
+  <center>
+    <img src="40va-7.gif" width="360" height="470" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The azimuth plot overlays two patterns, one in each direction. One of the key advantages of loading the reflector is that--with very little effort--we can reverse the beam's direction. The reflector called for a 55-Ohm load to achieve the desired pattern. Instead of using an inductor as the loading element, we may use a shorted length of feedline, in this case, 50-Ohm cable to match the feedpoint impedance of the driver element. The electrical length would be about 18.24', but the physical length will be the electrical length times the cable's velocity factor. Even a solid dielectric cable will yield a physical length of at least 12', so that cable running from each element can meet in the center of the 21' element spacing. Then, with a suitable remote switch, we can short one cable to make the inductively reactive load for the reflector and connect the other line to the main feedline. With a flip of the switch, we have reversed the beam's direction.</p>
+  <p>When we first encounter vertically polarized parasitic arrays, we are sometimes surprised by the facts that their gain is lower and their beamwidth is much greater than the patterns that we find in horizontal arrays using the same number of elements. The beam's gain value is about 3.3 dB higher than for a single T-cap dipole, but less than the bi-directional gain of the phase-fed pair of verticals. The beamwidth is nearly 140 degrees, providing very good coverage in each of the beams two possible directions. Although ground losses have a role to play in setting the gain of all vertical antenna types that are close to the ground, the vertical array's gain would not catch up to the gain of a horizontal counterpart until we reached a height above 20 wavelengths. The key factor in the gain and beamwidth difference that we get from rotating the beam 90 degrees along its real or virtual boom is a function of geometry. The arrangement of element tips restricts the beamwidth in the E-plane, that is, in the plane of the elements. However, the H-plane has no such restrictive influence. Indeed, in free-space, H-plane patterns of a 2-element parasitic array look very much like the pattern of a single vertical dipole but displaced in the direction of the forward gain. A Yagi must have many elements (and be very long) before the H-plane beamwidth narrows significantly.</p>
+  <p>Nevertheless, the reversible 2-element vertical array with T-cap elements provides good service across the 40-meter band. As the 50-Ohm SWR curve at the bottom of <b>Fig. 7</b> shows, the array will cover most of the band with less than 2:1 SWR. However, we early on noted that our bandwidth concerns covered more than just the SWR. We are also interested in how well the antenna performs in terms of gain and front-to-back ratio. <b>Fig. 8</b> provides a partial answer to our questions.</p>
+  <center>
+    <img src="40va-8.gif" width="744" height="396" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The gain curve in the figure shows a 0.4-dB range of forward gain value across the band, a value that we may consider fairly stable for such a wide amateur band using elements composed of relatively thin wire. In contrast, the front-to-back ratio remains above 10 dB only for a small portion of the band. However, it never drops below 5 dB. In fact, with careful adjustment of the reflector load, we may be able to center better the front-to-back curve for relatively similar performance at the band edges. A slightly higher load reactance--meaning a slightly longer length for the shorted stub--would likely do the job. Alternatively, we may favor either the CW-digital end of the band or the phone end of the band, according to our operating needs.</p>
+  <p>One limitation of the 2-element driver-reflector array is that we leave some portions of the horizon only weakly covered. It might be useful if we could cover the entire horizon with less than a 2-dB drop in gain around the 360-degree span--and at the same time maintain at least the forward gain that we obtained from the reversible 2-element array. To achieve that goal, we may think in triangles.</p>
+  <p>Suppose that we set up a an equilateral triangle of T-cap vertical dipoles. We may designate the corner containing the driven element as the apex. The remaining corner will contain identical T-cap dipoles, but loaded to form reflectors. <b>Table 5</b> provides the dimensions and test-frequency performance data, while <b>Fig. 9</b> supplies a sketch and the radiation patterns.</p>
+  <center>
+    <img src="40va-t5.gif" width="485" height="266" border="2" vspace="5" hspace="9">
+  </center>
+  <center>
+    <img src="40va-9.gif" width="400" height="470" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The T-legs are each 9.5' long for the individual dipoles. Each triangle side is about 30.6' long, yielding a distance between the apex and the center of the triangle base of 26.5'. It does not matter how we orient the T-legs of the dipoles, so long as the legs form a straight line for each dipole. In the triangle shown--and other dimensions are also possible--each reflector dipole requires a j90-Ohm load. We shall again use shorted transmission-line stubs as the source of the required inductive reactance. Since the driver impedance is close to 70 Ohms, we shall use 70-Ohm lines, which require a 19.9' electrical length, with physical shortening that depends on the VF of the line used in the assembly. We shall bring the stubs to a center point within the triangle for switching. At any time, one of the stubs will actually be an extension of the main feedline to the driver, while the switch to form the required reflector loads shorts the other two lines.</p>
+  <p>The system gain is about 3.2 dBi, about 0.5-dB higher than for a standard 2-element parasitic array and about 3.7-dB higher than a single T-cap vertical dipole. The 15-dB front-to-back ratio is also several dB higher than we found for the 2-element parasitic beam. With its higher gain, the triangle shows a beamwidth that is about 5 degrees narrower than the beamwidth of the simpler beam. Perhaps the key advantage of the triangle is its ability to cover the entire horizon with only a 2-dB gain deficit at the overlap points between the forward lobes of the beam in each of its positions. Although the switching may be more complex for the triangle than for the reversible beam, the electronics are simple and cheaper than a rotator.</p>
+  <p>The lower portion of <b>Fig. 9</b> shows the 50-Ohm and 70-Ohm SWR curves for the array. A 50-Ohm feedline from the switch junction to the equipment will be satisfactory if SWR values close to 2:1 are satisfactory at the band edges. However, a 70-Ohm line will reduce the maximum SWR value to about 1.5:1 at the band edges. With respect to the gain and front-to-back performance across the 40-meter band, <b>Fig. 10</b> supplies the appropriate sweep data.</p>
+  <center>
+    <img src="40va-10.gif" width="744" height="396" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The performance bandwidth of the triangle is generally similar to the performance bandwidth of the reversible array, but with a smaller range of value change across the band. The gain varies by only about 0.2 dB. The front-to-back ratio remains at 10 dB or more for the entire band. The band-edge values may be equalized by slight adjustments to the load values, that is, the length of the shorted stubs. With the smaller range of performance change over the 300 kHz of amateur allocation (in the U.S.), one may design a triangle for whole-band use and expect only small deficits at each band edge.</p>
+  <p><b>A 2-Element Phased Beam Using T-Cap Dipoles</b></p>
+  <p>Obtaining the maximum possible front-to-back ratio from 2 elements is difficult if not impossible using parasitic techniques and 2 vertical elements that are close to the ground. However, it is easily possible to improve the front-to-back performance and to obtain a nearly cardioidal pattern by phasing 2 T-cap dipoles in an endfire arrangement. In fact, we may do so using commonly available feedline materials, although we may have to do some experimentation to find the optimal cable for the task. Experimenting with modeling software is more rapid and less costly than experimenting with lengths of actual cable.</p>
+  <p><b>Table 6</b> profiles the phased array that uses two standard T-cap dipoles with 9.6' T-legs and a spacing of 21' (the same spacing used for the reversible parasitic array). Both dipoles use identical construction. The phase line uses two separate lengths that reach a junction where we shall connect the main feedline. The section to elements 1 in <b>Fig. 11</b> is 084' of RG62, 93-Ohm line with a VF of 0.84. Hence, the electrical length is 1'. The line does NOT undergo a half twist or reversal. However, the line from the junction to the rear element--from the same material--does undergo a reversal. It is 21' physically or 25' electrically.</p>
+  <center>
+    <img src="40va-t6.gif" width="549" height="284" border="2" vspace="5" hspace="9">
+  </center>
+  <center>
+    <img src="40va-11.gif" width="400" height="470" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The forward gain of the array is comparable to the gain of the reversible parasitic array. However, the front-to-back ratio climbs to over 26 dB at the design frequency. As well, the beamwidth increases to about 145 degrees, providing wide coverage in this essentially mono-directional array. The rearward quadrants of this array promise to be exceptionally quiet.</p>
+  <p>The cost of this type of performance lies in the odd feedpoint position before we add matching components. The low impedance has a high inductive reactance. However, the values are nearly optimal for a beta match, so long as we use an open or capacitively reactive stub across the feedpoint terminals. 14.3' of 50-Ohm VF 0.78 cable or its equivalent provides the necessary shunt component for the beta L-network that yields an impedance close to 50 Ohms. With the beta component in place, the SWR curve in <b>Fig. 11</b> shows well under a 2:1 SWR value at the band edges.</p>
+  <center>
+    <img src="40va-12.gif" width="744" height="396" border="2" vspace="5" hspace="9">
+  </center>
+  <p><b>Fig. 12</b> provides sweep data for the phased array relative to forward gain and front-to-back ratio. The gain rises steadily across the band over a 0.4-dB range. The front-to-back ratio peaks near the middle of the band and decreases to about 15 dB at the band edges. Compared to parasitic arrays, the whole-band front-to-back performance of the phased array rates very highly. Perhaps the major drawback to phasing is that the array does not succumb easily, if at all, to reversing direction.</p>
+  <p><b>Screen Reflectors and the T-Cap Driver</b></p>
+  <p>Screen or planar reflectors have been used since the 1920s in major arrays. Indeed, in their early years, some called them billboard reflectors. A solid surface properly sized will reflect radio waves in ways that are analogous to the reflection of light from a flat mirror. We find them in wide use in dipole arrays for short-wave broadcasting and in a wide variety of UHF antennas. More recently, I have recommended their use beneath a number of NVIS antennas to increase gain, essentially by elevating the quality of the reflective plane beneath the antenna. In fact, the wire-grid ground improvement technique noted earlier represents a different application of the same technology.</p>
+  <p>A wide range of experience shows that planar or screen reflectors are most effective in improving directional gain when they exceed the dimensions of the driver element by between 0.45 and 0.55 wavelength both horizontally and vertically. We can achieve this goal horizontally with our T-cap vertical dipole driver, but any screen reflector will be vertically challenged. The ground, of course, is one limit, preventing the reflector from extending below the driver by the desirable amount. Vertically, we shall be limited as well by some of the height restrictions that we set for this project. If we limit ourselves to a 50' top height, the forward gain will be only a little greater than the gain of the reversible beam. A top height of 100' yields perhaps a half-dB more gain than a 70' height. Therefore, for the comparisons that we shall show, the height of the screens will run from just above ground level to 70'. One might hang such a screen between two widely separated towers that support antennas for higher amateur bands.</p>
+  <p>The optimum width turns out to be just about 140', which is 1 wavelength at the test frequency (7.15 MHz) and a half wavelength on each side of the driving dipole. Screen reflectors do not have to be solid surfaces to act like solid surfaces in the HF range. Chicken-wire fencing and open-weave materials will appear solid to 40-meter energy. The model for the screen-reflector array uses a wire-grid, as shown in <b>Fig. 15</b>. The dimensions and performance data appear in <b>Table 7</b>.</p>
+  <center>
+    <img src="40va-t7.gif" width="613" height="192" border="2" vspace="5" hspace="9">
+  </center>
+  <center>
+    <img src="40va-13.gif" width="360" height="470" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The selected distance from the driver to the reflector is 35' for an array using a single T-cap dipole driver and a 140' by 70' screen reflector. You may ground the screen bottom wires with no effect on performance, but with a considerable affect on safety. The driver uses 9.66' T-leg lengths for an 80-Ohm resonant impedance. With a planar reflector, you may vary two items to set a feedpoint impedance: the dipole dimensions and the spacing from the reflector. In general, closer spacing produces lower feedpoint impedance levels, but narrower operating bandwidths. Each space adjustment will change the coupling between the reflector surface and the driver, requiring adjustments to the T-leg lengths to return to resonance.</p>
+  <p>The chief merits of the planar reflector array are forward gain and operating bandwidth, when we compare the results to the reversible parasitic array. The top height of the reflector limits the front-to-back ratio to about 13 dB. A top height of 100' would have added another dB to the ratio, while a more ideal (and unrealistic) height of 140' would add a further dB or two. The array's forward gain is close to 4.7 dBi, about 1.8 dB higher than the 2-element parasitic array. The forward gain comes at the expense of the beamwidth, which is down to 83 degrees, about 50 degrees narrower than the beamwidth of the parasitic beam.</p>
+  <p>Despite the 80-Ohm resonant feedpoint impedance at the design frequency, the 50-Ohm SWR curve in <b>Fig. 13</b> reveals a very low rate of impedance change across the 40-meter band. That curve shows less than 2:1 SWR at the band edges. The lower 80-Ohm curve shows less than 1.5:1 SWR at the band's upper and lower limits. The remaining prime operating parameters are equally slow to change, as shown in the sweep data in <b>Fig. 14</b>.</p>
+  <center>
+    <img src="40va-14.gif" width="744" height="396" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The gain across 40 meters changes by only about 0.1 dB, while the front-to-back ratio varies by just over 0.5 dB. It is possible to design driver elements with an inherently wider bandwidth and to use the array to cover both 40 and 30 meters with very little change in performance.</p>
+  <p>One common method of constructing a screen is to use a series of wires polarized as the driver element. To sample that option, I reconstructed the screen reflector using 29 AWG #12 wires at 5.0' intervals. The overall horizontal and vertical screen reflector dimensions remained the same. Doubling the wire size (to AWG #6) yielded no performance improvements. The only other change relative to the wire-grid screen was a 3' increase in the spacing of the driver from the reflector, as shown in the dimensions in <b>Table 8</b>.</p>
+  <center>
+    <img src="40va-t8.gif" width="613" height="205" border="2" vspace="5" hspace="9">
+  </center>
+  <center>
+    <img src="40va-15.gif" width="360" height="470" border="2" vspace="5" hspace="9">
+  </center>
+  <p>As the data and the patterns in <b>Fig. 15</b> show, the vertical-wire reflector is not quite as effective as the wire-grid version. Gain drops by about a half dB, while the front-to-back ratio decreases by well over 2 dB. Both decreases are indications that the wire version of the screen requires a taller top height or is perhaps "leakier" than the wire-grid. As well, the resonant impedance is 5 Ohms higher and does not produce 50-Ohm SWR values under 2:1 without further matching. However, the lower 75-Ohm SWR curve in the graph does track the wire-grid's 80-Ohm SWR curve very well.</p>
+  <center>
+    <img src="40va-16.gif" width="744" height="396" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The sweep information in <b>Fig. 16</b> shows the same broad curves for both the forward gain and the front-to-back ratio. The gain range is about 0.15 dB across the band, while the front-to-back ratio changes by only 0.8 dB. Even with somewhat lesser performance than the wire-grid reflector, the vertical-wire screen still enjoys a considerable advantage over a parasitic reflector with respect to broadband characteristics.</p>
+  <p>To compensate, even if only partially, for ground losses, one may tilt the reflector back, as viewed from bottom to top. Using an 11' tilt (35' at the bottom and 46' at the top for the distance between the driver and the vertical-wire reflector), it is possible to add a few tenths of a dB to the forward gain and a similar amount to the front-to-back ratio. However, the exercise has its own consequences, as revealed in the overlaid patterns of <b>Fig. 17</b>.</p>
+  <center>
+    <img src="40va-17.gif" width="453" height="175" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The peak gain lines both forward and rearward are virtually indistinguishable in the patterns. However, note the increased high angle radiation, especially in the rearward lobes. As well, radiation directly overhead to the driver has also increased. In the end, a vertically oriented screen appears to yield the best combination of performance and patterns.</p>
+  <p>Earlier in these notes, I promised a second look at the in-phase-fed pair of T-cap dipoles spaced 1/2 wavelength apart. The dipole pair produced a bi-directional pattern with a maximum gain that was over 4 dB stronger than a single T-cap dipole. Since we can readily feed the pair of dipoles from a single feedpoint, using a pair of equal-length lines, we might wonder what would happen if we place the broadside array in front of a screen reflector.</p>
+  <p>Ideally, the screen should be over 200' wide. Understanding that we lose performance as we shrink a screen below its ideal proportions, let's continue to use the smaller 70' by 140' wire-grid screen. With the screen 40' behind the driver, the T-legs of each dipole are 10.16' long, longer than for a single dipole driver, but slightly shorter than the T-legs on the bi-directional array. As shown by the dimensions in <b>Table 9</b> and the outline sketch in <b>Fig. 18</b>, we lose a good bit of the horizontal screen extension that optimizes performance. Nevertheless, the performance is noteworthy.</p>
+  <center>
+    <img src="40va-t9.gif" width="455" height="264" border="2" vspace="5" hspace="9">
+  </center>
+  <center>
+    <img src="40va-18.gif" width="360" height="470" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The 6.95-dBi forward gain value for the array may require some perspective. First, the value is almost 2.3-dB higher than the gain we obtain from a single driver and the same reflector. A sign that the screen is less than optimal in size comes from the fact that the screenless bi-directional array produces a gain value that was 4 dB greater than a single T-cap dipole. Nevertheless, the phased dipole array and screen produce almost 7.5-dB gain relative to a single T-cap omni-directional dipole. The forward beamwidth is down to 58 degrees, dictating careful aiming of the array.</p>
+  <p>The front-to-back ratio is not outstanding, at about 11.4 dB, another sign of using a small screen reflector. Because the unmatched feedpoint impedance of each dipole is in the vicinity of 90 Ohms, we may run equal lengths of RG-62 (VF 0.84) to a center point between the dipoles and obtain a net impedance of 48.9 Ohms. As the 50-Ohm SWR curve shows, the array easily covers 40 meters with a maximum SWR that is less than 1.5:1.</p>
+  <center>
+    <img src="40va-19.gif" width="744" height="396" border="2" vspace="5" hspace="9">
+  </center>
+  <p>As the sweep data in <b>Fig. 19</b> reveal, the dual-dipole-driver array is as stable across 40 meters as the other sample screen-reflector array. Both the gain and the front-to-back ratio show changes of just over 0.2 dB across the band.</p>
+  <p>The construction of a screen-reflector is the most difficult portion of the overall project. Therefore, one may well wish to place drivers on either side of the screen for reversible service using separate feedlines all of the way to the equipment location. Whichever driver is in use (whether single or double), the inactive one will remain invisible due to the screen's reflection characteristics. Screen reflector arrays may not be for everyone, but they may serve a few 40-meter operators.</p>
+  <p><b>Conclusion</b></p>
+  <p>The reason for our title is now clear. We have not only surveyed comparatively the performance of vertical antennas and arrays on 40 meters, but we have added screen reflector arrays to the list that we normally see. This last group of arrays presents serious construction challenges, but offers in return increased gain over the values that we may obtain from parasitic arrays.</p>
+  <p>All of the driving elements in our comparisons have used AWG #12 copper wire in a T-cap arrangement that extends from 5' to 40.5' above average ground. We changed the length of the T-legs to obtain a resonant feedpoint impedance. The T-cap dipoles provide a uniform element length to help validate the comparisons. Although scarcely longer than 1/4 wavelength, the T-cap dipoles lose very little performance relative to the reference full-length wire dipole with which we began these notes.</p>
+  <p>Just because we cannot obtain the gain level of a phase-fed dipole pair with a screen reflector does not relegate the intermediate designs to uselessness. Indeed, many operators use T-cap and similar 40-meter vertical antennas with surprisingly good results. The parasitic arrays offer reversibility and even--with the triangle--full horizonal coverage at the flip of a switch. For maximum front-to-back ratio, the phased array is difficult to surpass, even though it uses no complex networks to achieve a nearly cardioidal pattern.</p>
+  <p>By setting the entire range of vertical arrays on common ground at a common frequency with common construction, you may gather a more precise sense of the benefits and costs as you increase the complexity of a vertical array. Virtually all of the designs will scale directly to 30 meters, although bandwidth there is not a major concern beyond the realm of allowing rather casual construction without incurring performance losses. Scaling to 80 meters is also possible, although for most amateurs, the resulting element sizes may prove to be prohibitive. A T-cap dipole of the present design will extend from about 10' to about 81' at the top, depending on the precise frequency to which one scales the design. The T-legs can be trimmed for resonance if you stick with the AWG #12 wire rather than doubling its diameter. If the amount of trimming is not too great, adjusting only the lower T-legs will not disturb the centering of the feedpoint to any significant degree. (The presence of ground below the bottom of the antenna and essentially free space above the top already disturbs the balance that we presume when we physically center the feedpoint on a vertical dipole. Common-mode current attenuation devices are necessary adjuncts to any of these arrays.)</p>
+  <p>These notes have focused on performance comparisons and provided very few construction notes. Building any of these arrays, from the simplest to the most complex, will be an exercise in making use of available supports and in using what nature or prior antenna construction provides. In virtually all cases, the result will be far less expensive than a strong tower able to hold the weight of a horizontal beam that shows higher-angle elevation lobes and questionable operating bandwidth.</p>
+  <p><i>First printed in</i> QRP Quarterly<i>, January, 1997. Updated 01-22-97. &COPY; L. B. Cebik, W4RNL. Data may be used for personal purposes, but may not be reproduced for publication in print or any other medium without permission of the author.</i></p>
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+  <h1 style="text-align: center;">80-Meter Dipoles and Inverted-Vs<br>
+  A Graphical Scrapbook</h1>
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+  <p style="text-align: center; font-weight: bold;"><a href="80dp.pdf">80-Meter Dipoles and Inverted-Vs A Graphical Scrapbook</a></p>
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+<!DOCTYPE html>
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+  <meta name="keywords" content="HF Multi Band, Doublet, Matching">
+  <title>10 Frequency Asked Questions about the All-Band Doublet</title>
+  <link rel="stylesheet" href="../styles.css">
+</head>
+<body>
+  <center>
+    <h1><img src="../images/wirecad.gif" align="left" width="125" height="62" border="2" vspace="15" hspace="9"> <img src="../images/w4rnlwwr.gif" align="right" width="99" height="83" border="2" vspace="5" hspace="10"> 10 Frequency Asked Questions about the All-Band Doublet</h1>
+  </center>
+  <center>
+    <h3>L. B. Cebik, W4RNL</h3>
+  </center>
+  <center>
+    <img src="../images/colorbar.gif" align="middle" width="540" height="4">
+  </center>
+  <p>The all-band doublet horizontal wire antenna has a history almost as long as amateur radio itself. Despite all the words and diagrams in handbooks over the years, newcomers still send me questions about the antenna. I have collected the questions and boiled them down to 10, all of which have many variations. The goal in tackling these frequently asked questions is to help newer hams erect a successful antenna system.</p>
+  <p><b>1. What is an all-band doublet?</b> The all-band doublet is actually an antenna system and not just an antenna alone. <b>Fig. 1</b> shows the basic elements of the system. The horizontal center-fed wire forms the antenna proper, which accounts for the radiation of transmitted energy and the reception of incoming energy. The parallel transmission line transfers the energy from the antenna to the antenna tuner (or antenna-tuning unit, the ATU) or vice versa. We insert the tuner because the impedance that shows up at its terminals will vary widely from one band to another. So we need a way of matching the impedance at the tuner terminals to the standard 50-Ohm input and output impedance of the transceiver.</p>
+  <center>
+    <img src="abd-1.gif" width="507" height="231" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The antenna wire itself can have many lengths, but should be about &frac12; wavelength at the lowest operating frequency. <b>Table 1</b> shows common doublet lengths that have appeared in handbooks since the 1930s. It also shows the ham bands covered by the antenna. Note that the 100' wire, while somewhat shorter than &frac12; wavelength, can be pressed into service on 80 meters, and the 67' wire might be used on 60 meters. However, there are limits that we shall explore as we proceed through the questions.</p>
+  <center>
+    <img src="abd-t1.gif" width="438" height="125" border="2" vspace="5" hspace="9">
+  </center>
+  <p><b>2. What's the difference between a doublet and a dipole?</b> This interesting question has 2 answers: none and a lot. Conversationally, the term "dipole" often refers to any antenna that looks like a dipole, that is, a center-fed wire antenna with a feedline going to the shack. In this context, we also tend to call any end-fed antenna a Zepp (although there is a center-fed extended double Zepp) and to refer to any off-center-fed antenna as a Windom (although the original Windom had only a single feed wire).</p>
+  <p>In more precise terms, the coax-fed dipole that we sometimes set up for single-band use is a more complex antenna than its appearance suggests. It is actually a center-fed resonant &frac12; wavelength dipole. The center-feedpoint is obvious from the position of the feedline. It is resonant since the feedpoint impedance is (almost) purely resistive, with little or no reactance. The length is electrically &frac12; wavelength, which for any real wire or tubular element turns out to be shorter than a physical half wavelength. Finally, it is a dipole because, as <b>Fig. 2</b> shows, the charge is minimum at the center feedpoint and maximum at the element ends. As a result, the current is maximum at the center and minimum at the wire ends. The dipole undergoes only one transition in charge and in current from the center to the wire end.<sup>1</sup></p>
+  <center>
+    <img src="abd-2.gif" width="379" height="274" border="2" vspace="5" hspace="9">
+  </center>
+  <p>When we use the antenna on many bands, it becomes electrically longer, because the length of a wave grows shorter with rising frequency. Hence, the charge and current patterns do not satisfy the dipole conditions above the lowest band or two. <b>Fig. 3</b> shows the current distribution along a 135' wire at 80 meters and at 10 meters. Since the current does not follow the dipole pattern, the charge density is also different from a dipole. In this case, there are many transitions and the current is not maximum at the center feedpoint.</p>
+  <center>
+    <img src="abd-3.gif" width="593" height="266" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The term doublet is more generic and less fully descriptive than others. However, it also has a history. In the 1930s, it served as a label for a center fed wire with a special feed system. Later, the antenna was renamed the delta feed and the term doublet became a generic term for center-fed antennas of any length.<sup>2</sup> Hence, our antenna is an all-band doublet.</p>
+  <p><b>3. Do I need to measure the wire for precise resonance on the lowest band?</b> In a word, no. When we set up a resonant monoband dipole, we want it to achieve resonance or the lowest possible SWR with our coax cable feedline. However, the all-band doublet antenna system uses (normally) high impedance parallel transmission line. Small variations in antenna wire length will make no difference to doublet performance or to our ability to match the impedance at the shack end of the feedline. We sometimes see radiation patterns for a Yagi antenna change shape as we move from one end of an amateur band to the other. However, the patterns produced by the all band doublet change very slowly with frequency. For example, if we only have room for 125' of wire, then it will do very well and yield the same results as a 135' wire on the lower HF bands. On the upper HF bands, we might see some change in the feedpoint impedances between the two wire lengths, but they normally will not be severe and certainly not large enough for us to abandon the antenna. The recommended lengths in <b>Table 1</b> are ballpark figures, not precise lengths.</p>
+  <p><b>4. Why do I need parallel transmission line? Why not just coax? Or coax with a 4:1 balun?</b> To get our hands on this question, let's consider only one of the possible doublet lengths: 135'. For this version of the doublet, we can look at the numbers in <b>Table 2</b>. The second column lists the approximate feedpoint impedances for each HF amateur band. These numbers will vary with the exact length of the wire and the height above ground. However, the approximations will serve well for our demonstration.</p>
+  <center>
+    <img src="abd-t2.gif" width="601" height="245" border="2" vspace="5" hspace="9">
+  </center>
+  <p>Suppose that we connect a typical coaxial cable to the feedpoint and use 100' of the line to reach the shack. RG-8X is popular these days because it is light and easy to handle. How much energy will we lose if we use this cable as a feedline? We can arrive at some answers by using a program like TLW. This highly useful software, written by Dean Straw, N6BV, accompanies <i>The ARRL Antenna Book</i>, which is a worthy long-term investment for any ham. In the table, columns 3 and 4 show the 50-Ohm SWR for each of the impedances and the total cable losses. Notice how many of the loss entries exceed 10 dB. With a 10-dB loss, only 1/10 of the energy at one end of the line is available for use at the other end of the line. The reduction applies whether we are transmitting or receiving.</p>
+  <p>The last 2 columns show the SWR for a 450-Ohm parallel transmission line. The type specified uses a vinyl coating with windows along the way. The vinyl coating is simply a good way to keep the wires evenly spaced, but it does introduce losses that are slightly greater than open or true ladder line (bare wire with periodic spacers). Note that even with the highest SWR levels, losses do not exceed 0.5 dB or a little over 10% of the power, even with 100' of the line.</p>
+  <center>
+    <img src="abd-4.gif" width="722" height="481" border="2" vspace="5" hspace="9">
+  </center>
+  <p>Parallel lines do have limits however. Remember that we recommended &frac12; wavelength at the lowest frequency as the shortest antenna wire length. We also suggested that we might press shorter wires into service, but we did not say how much shorter. Let's see what happens below 80 meters as we shorten the wire from a &frac12; wavelength starting point. <b>Fig. 4</b> shows the approximate resistance and reactance. Although the curves appear to track each other, remember that the downward path of reactance actually represents increasing capacitive reactance. As we shorten the doublet or lower the frequency, the feedpoint resistance decreases steadily, while the capacitive reactance increases steadily. The result will be a very high 450-Ohm SWR on the parallel line. It will rise to the point where even the seemingly low-loss line shows significant power losses along the way. As a practical matter, try to keep the antenna at least 3/8 wavelength or longer at the lowest frequency used if you cannot manage &frac12; wavelength. Remember that you can always zigzag the wire legs or let the ends droop downward (but always with their ends out of human reach) in order to lengthen the wire to the full &frac12; wavelength at the lowest frequency.</p>
+  <p><b>5. What's the most important factor in setting up an all-band doublet? Or, we put up a low-band doublet for Field Day about 10-15' off the ground. We did not make many contacts? What was wrong?</b> The question's second form gives us the answer to the general question. With an all-band doublet, there is no substitute for height. However, hams must work with real conditions and not ideals.</p>
+  <p>Let's continue to use the 135' doublet as our antenna and see what happens at various antenna heights that hams actually use. 20' is a typical Field Day height for wire antennas due to the difficulty of erecting and sustaining higher supports. 40' is a nice round number for a backyard doublet supported by mature trees. 60' is out of reach for amateurs unless they have a tower or two supporting rotatable beams. Now look at <b>Table 3</b>. It lists for each sample operating frequency the height above ground as a fraction of a wavelength.</p>
+  <center>
+    <img src="abd-t3.gif" width="451" height="246" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The height above ground when measured in terms of a wavelength is the most important factor that determines the elevation angle of a horizontal antenna's radiation. (Remember that the radiation angle is also the angle of reception sensitivity.) <b>Fig. 5</b> provides a catalog of typical elevation patterns for the doublet. Each pattern uses the headings for maximum gain as a basis. The missing bands would show elevation angles of maximum radiation that are part way between the bands in the illustration.</p>
+  <center>
+    <img src="abd-5.gif" width="600" height="500" border="2" vspace="5" hspace="9">
+  </center>
+  <p>Note that at 80 meters, all three heights are so low that we detect very little elevation pattern difference. The pattern begins to change significantly as we raise the antenna to 60' when operating on 40 meters. The 20-meter pattern becomes very usable for low-angle skip radiation when we raise the wire to 40', a little bit more than &frac12; wavelength. 20' is a little over &frac12; wavelength on 10 meters, and so we obtain reasonable basic performance on that band.</p>
+  <p>These notes and graphics cannot change your backyard or field conditions. However, they do provide food for thought. For example, if you really want to operate on 80 and 40 meters, but cannot get the horizontal antenna high enough as a fraction of a wavelength, then you may wish to consider alternative antennas. You might achieve better performance on the lowest HF bands with a different wire antenna, such as the inverted-L.<sup>3</sup></p>
+  <p><b>6. I carefully set up my 135' doublet to be broadside to Europe. However, on 15 and 10 meters, signals are much stronger to Africa than to Europe. Is it propagation?</b> Although propagation affects all HF communications, the most likely source of the weak European signals is a misunderstanding of the azimuth patterns produced on the various amateur bands by the 135' doublet.</p>
+  <center>
+    <img src="abd-t4.gif" width="631" height="250" border="2" vspace="5" hspace="9">
+  </center>
+  <p>As a center-fed wire antenna grows longer in wavelengths, the number of lobes that it produces increases. <b>Table 4</b> lists the length of our 135' doublet as a function of a wavelength on each operating frequency. It also lists the number of lobes produced in the azimuth pattern, that is, around the horizon. The numbers may seem odd, but there is nothing disorderly about them. As the antenna grows longer (or we increase the operating frequency), lobes emerge, reach a peak value, and then disappear--to be replaced by other emerging lobes. Three main points describe the process. Let n be the length of the antenna rounded to full wavelengths.</p>
+  <blockquote>
+    1. If the antenna is n wavelengths long, then the number of lobes will be 2n. On 40 meters, the antenna is about 1 wavelength, so there will be 2 lobes. At 10 meters, the antenna is about 4 wavelengths, so we shall find 8 lobes.
+    <p>2. If the antenna is n.5 wavelengths, we shall find the lobes for n wavelengths and the lobes for n+1 wavelengths at close to equal strength. So add the number of lobes for n wavelengths and the number of lobes for n+1 wavelengths to arrive at the total number of lobes. At 17 meters, the antenna is about 2.5 wavelengths. A 2 wavelength wire gives us 4 lobes and a 3 wavelength wire yields 6 lobes. So at the intermediate length, we shall find 4+6=10 lobes.</p>
+    <center>
+      <img src="abd-t5.gif" width="584" height="256" border="2" vspace="5" hspace="9">
+    </center>
+    <p>3. When the wire is close to n or to n.5 wavelengths, the strongest lobes will be those farthest from broadside to the wire, that is, closest to in line with the wire. <b>Table 5</b> provides the modeled performance data of the 135' doublet at a height of 40' above ground. <b>Fig. 6</b> translates those numbers into a gallery of azimuth patterns. The virtual antenna runs up and down on the graph page. Because the take-off (TO) angle (or the elevation angle of maximum radiation) is so high for 80 through 40 meters, the azimuth patterns use an arbitrary elevation angle of 45&deg;. All other patterns use the actual TO angle.</p>
+  </blockquote>
+  <center>
+    <img src="abd-6.gif" width="600" height="650" border="2" vspace="5" hspace="9">
+  </center>
+  <p>Except for 30 meters, where the null between the inner lobes is hard to detect, all of the patterns clearly exhibit the number of lobes calculated in <b>Table 4</b>. Since all of the lengths are close to either a full wavelength or the half wavelength mark between full wavelengths, the strongest lobes are those nearest to being in line with the wire. (When the wire is close to n.25 or n.75 wavelengths, other lobes may dominate.)</p>
+  <p>Note that when the length is n.5 wavelengths, the large number of lobes in the pattern forces the strongest lobes to be closer to in line with the wire than for the next whole number of wavelengths. Hence, the angle of the lobes away from broadside is greater on 17 meters than on 15 meters--and greater on 12 meters than on 10 meters. Also note that the larger the number of lobes in a pattern, the narrower the beamwidth of each lobe.</p>
+  <p>If we had chosen a 67' doublet, the antenna would be &frac12; wavelengths on 40 meters, 1 wavelength on 20 meters, and 2 wavelengths on 10 meters. Since the azimuth lobes are functions of the wire length in wavelengths, we would obtain different lobe patterns than for the 135' wire. In fact, the 67' wire pattern on 40 would resemble the 135' pattern on 80, and the 67' 10-meter pattern would look very much like the 135' 20-meter pattern.</p>
+  <p>How you orient a center-fed doublet depends on understanding both the elevation and the azimuth patterns for the wire. The azimuth patterns show where your signal is likely to go, while the elevation patterns tell you whether the energy is likely to fall within the skip zone. Orient the doublet so that the pattern for the most used band (or bands) covers your most desired target(s) with a strong, low-angle lobe.</p>
+  <p>So far, we have concentrated on the wire or antenna-proper portion of the all-band doublet antenna system. We briefly explored the main reason for needing to use parallel transmission lines to connect the antenna to the antenna tuner. Hams who are used to using coax often ask a number of other questions about parallel feedlines.</p>
+  <p><b>7. Can I run the parallel feeders in a PVC tube underground or under my house?</b> This question is actually a confession by the newcomer that he or she knows how to handle coaxial cable, but not parallel feedline. In a coaxial cable, the energy fields exist between the outer surface of the inner or center conductor and the inner surface of the outer conductor, also called the braid. Hence, if the cable has an outer jacket that can handle soil, burying it does not affect its use or operation. As well, we can run the cable near to other wires without significant difficulty.</p>
+  <p>Parallel feedline is also called open-wire transmission line, and for good reason. Regardless of whether the wires have insulation, they are open in the sense of having fields that are not confined by the structure. Although the main portion of the field is between the two wires, it also extends around the pair of wires for a considerable distance--up to a few times the spacing between the wires. Nearby conductive and semi-conductive materials can disturb the balance between the lines and cause them to radiate--a job we want the antenna proper to do. As well, we may lose some energy to those objects. So, in a nutshell, the answer to the question is no. Do not run the transmission lines close to or within the ground, even if you give them the double insulation of a conduit.</p>
+  <p>Commercially available lines come in 3 general types, each with a different characteristic impedance, construction, velocity factor (VF), and loss value. 300-Ohm transmitting twinlead, sometimes flat and sometimes tubular, has a VF of about 0.80 and a loss of about 0.17 dB per 100' at 3.5 MHz. Remember that line losses increase with frequency. 450-Ohm window line, a form of flat twinlead with cutouts to minimize the vinyl between the wires, has a velocity factor of about 0.91 and a 0.07-dB loss per 100' at 3.5 MHz, half the loss of 300-Ohm line. 600-Ohm open-wire ladder line typically has a velocity factor of about 0.92 or higher and a loss of only about 0.03 dB per 100' at 3.5 MHz. There are also commercially available ladder lines in the 400-500 Ohm range, and their VF and loss values would resemble those of the 600-Ohm line. Of the 3 types, 450-Ohm window line is perhaps the most popular for all-band doublets.</p>
+  <p>Parallel feedline has a few simple rules for effective placement to maximize energy transfer from the tuner to the antenna proper. Keep line runs as straight and in the clear as possible. Straight, clear runs are as important indoors as outdoors. Straight is self-evident. Wherever possible, keep direction changes shallow. Never let the line fold back upon itself or roll it in a coil. Clear means as far from other objects as possible, and in no case less than several times the line spacing away from anything.</p>
+  <p>Of course, we must bring the line indoors. We can use a short through-wall PVC pipe, perhaps with caps that have slots to keep the line centered. Or we can use a wood or plastic plate with feed-through insulators. The difference in spacing and bolt size on the board relative to the line is not important: it may create a small impedance bump but will minimize losses. Outdoor supports can be of two general types: rings or clamps. We can suspend non-conductive rings (slices of PVC or similar) from limbs and posts to support the line on its way to the antenna. As well, we can create non-conductive guides or clamps that extend outward from tree trunks, posts, or walls to route the transmission line. Be sure to use enough supports.</p>
+  <p>At the junction with the antenna, use a strain-relief fixture. A simple insulator may keep the line from being pulled by the antenna wire. However, over a relative short time, the feedline wires will flex back and forth until they break. A fixture that minimizes the flexing at the junction itself will make the connections much more durable.</p>
+  <p><b>8. Will the feedpoint impedance in the tables appear at the antenna tuner terminals?</b> If the feedline is precisely a multiple of an electrical half wavelength, then the feedpoint impedance will reappear at the far end of the line. (The other condition that would allow the feedpoint impedance to reappear is an exact match between the feedpoint impedance and the characteristic impedance of the cable. With 450-Ohm line, <b>Table 2</b> makes it clear that this condition will not exist.) When the characteristic impedance of the line does not match the feedpoint impedance, the line becomes a continuous impedance transformer and shows a different impedance at each step between the feedpoint and each half wavelength or 180&deg; point along the line.</p>
+  <center>
+    <img src="abd-7.gif" width="722" height="481" border="2" vspace="5" hspace="9">
+  </center>
+  <p><b>Fig. 7</b> shows one example of the transformation and applies to 450-Ohm transmission line and a feedpoint impedance of 2000 - j2000 Ohms. This impedance is similar to some values in <b>Table 2</b>. If the reactance had been inductive instead of capacitive, we would see similar curves, but the peaks would appear at 10-15&deg; position along the line (where 0&deg; is the antenna feedpoint and 180&deg; is a half wavelength down the line). Note the very low resistance and the relatively low reactance that appear over much of the line's length. For this reason, placing a 4:1 balun in the line may be a poor choice for converting the balanced line to a single-ended or unbalanced line. The transformer may end up converting a low impedance to a very low impedance, regardless of the balun's ability to handle the reactance at its terminals.</p>
+  <center>
+    <img src="abd-8.gif" width="722" height="481" border="2" vspace="5" hspace="9">
+  </center>
+  <p>Let's change the feedpoint impedance to 20 + j100 Ohms. Some impedances for very short doublets (less than about 3/8 wavelength) show a higher reactance (capacitive), but the peaks become very high and the resistance becomes very low--a very difficult situation to graph. <b>Fig. 8</b> graphs the resistance and reactance of the selected values along the line. Note that the peaks occur just before the 90&deg; or halfway position. If the reactance had been capacitive, the same peaks would appear just past the halfway point along each half wavelength of transmission line. Once more, note for how much of the line the resistance and reactance are low to very low.</p>
+  <p>The two samples show some of the extremes of impedance transformation along a 450-O transmission line. The closer the feedpoint impedance is to the line's characteristic impedance, the less radical will be the transformation of resistance and reactance. It pays to have a small calculation program to assist in finding and visualizing the data. Earlier, I mentioned the N6BV program, TLW. You can obtain similar graphs on it. In addition, the graphs will show the effects of line losses.</p>
+  <p>The graphs show an electrical half wavelength of line. The physical length of such a line will vary with the operating frequency. Hence, it is very difficult (although not impossible) to design a feedline system so that on each band we end up with just about the same impedance at the shack entry. Most amateurs let the antenna tuner do the work of transforming whatever impedance appears at the terminals to the transceiver's required 50 Ohms.</p>
+  <p><b>9. What kind of antenna tuner is best for an all-band doublet?</b> The best type of antenna tuner is one with a configuration that naturally has an unbalanced or single-ended input--to accept the transceiver's coaxial cable--and a balanced output. From the earliest days of amateur radio, a common tuner meeting these conditions has been the link-coupled tuner. <b>Fig. 9</b> provides a simplified schematic diagram of one version of this tuner. It received its name because the input side used a small coil or link that is inductively coupled to the tank or parallel tuned circuit on the output side. The most effective forms of this tuner used additional components on the output side to compensate for the reactance at the terminals. Taps at every turn (or at least at every other turn) of the tank coil allowed the user to find a setting that came closest to providing a good match and maximum power transfer at the same time. The link might also have switched taps with the later addition of the so-called variable coupling series capacitor. In fact, the series capacitor compensates for remnant reactance on the input side, allowing a purely resistive input impedance. Johnson Matchboxes, with simplified tank tapping, a fixed link, and no series input-side capacitor, became famous and still appear at hamfests.</p>
+  <center>
+    <img src="abd-9.gif" width="400" height="275" border="2" vspace="5" hspace="9">
+  </center>
+  <p>From the late 1960s onward, the single-ended network came to rule the commercial manufacture of antenna tuners. <b>Fig. 10</b> shows 4 popular configurations, with the CLC-T being the most common. It was perhaps the cheapest to produce in a period of rapidly rising component costs. It would also handle a very wide range of impedances at the output terminals. However, the CLC-T was a high-pass filter network and hence provided little harmonic suppression for older rigs. Like all of the single-ended configurations, it required a balun on the output to allow for balanced lines. The standard version of the balun used a 4:1 impedance ratio either though a misunderstanding of the impedances likely to be present at the terminals or because such baluns were cheaper to make than 1:1 baluns. The baluns were transmission-line transformers that are most efficient when the reactance is very low. Most balanced lines, however, did not meet this condition. The average operator did not have multiple tuners to compare and so remained unaware that on some bands with some tuners, efficient power transfer might not occur.</p>
+  <center>
+    <img src="abd-10.gif" width="510" height="400" border="2" vspace="5" hspace="9">
+  </center>
+  <p>In recent years, interest in antennas that require parallel transmission lines has surged, spurring the development of new inherently balanced tuners. <b>Fig. 11</b> shows three varieties that are either on the market or in handbooks. The single-ended CLC-T network is usable with special precautions not to ground any component except the transceiver side of the 1:1 input balun that is common to all of the tuner designs. One commercial tuner uses a balanced CLC-T network, but the most common balanced network tuner on the market is the reversible-L circuit. Versions exist for high power use. However, as the operating frequency increases, the range of impedances that the reversible-L will match with standard components grows more limited.</p>
+  <center>
+    <img src="abd-11.gif" width="475" height="450" border="2" vspace="5" hspace="9">
+  </center>
+  <p>If you will buy a tuner with an all-band doublet in mind, then one of the balanced network tuners may be the best bet. However, if you already have a tuner--even a single-ended network with a 4:1 balun on the output side--you might as well try it out. Since none of the tuners comes with a relative output indicator, you will have to estimate efficiency on each band indirectly. If you obtain a good match following the maker's suggestions for the best component settings, check the temperature of the balun after (not during) operation. If the balun is warm to the touch, it likely is converting some part of your transmitted energy into heat. In general, the broader the tuning, the lower the tuner losses, although there are exceptions to this rule of thumb. In a tuner designed for all of the HF ham bands, tuning will naturally become sharper with rising frequency.</p>
+  <p>If you cannot obtain a match on a given band, then try inserting a length of transmission line, preferably outdoors. Using knife switches, relays, or a simple manual changeover, add a few feet of line between the line ends of a break that you intentionally make in the feeders. Form the insertion into a single large loop to avoid unwanted self-coupling, and use standard precautions to prevent coupling to other objects. Since the transmission line is a continuous impedance transformer, the new values of resistance and reactance at the tuner terminals may fall within the tuner's range. Since every tuner has a limit to the range of resistance-reactance combinations that it will handle, the potential need for a revision in the total feedline length may apply to all antenna tuner designs.</p>
+  <p><b>10. My all-band doublet is 50' high and uses open-wire feeders. It works well, but I get a lot of RF interference at home, and my rig sometimes locks up in transmit on CW. How can I overcome these problems?</b> Unwanted coupling into home electronics and into the rig itself has almost as many causes as there are errors that we may make in installing parallel feedlines. The first step is to ensure that all station equipment is well grounded to an earth ground as close to the rig as may be feasible. The second step is to consider rearranging the station so that you position the antenna tuner at the place where the feedline enters the building or shack. Well-grounded coax braid is less likely to couple RF energy to other lines and objects than open-wire transmission line.</p>
+  <p>The third step is to check the routing of the transmission line as it approaches the entry point. Ideally, the line should approach the entry perpendicular to the wall or window. If the line runs vertically down a wall, it may couple energy into various power, telephone, or computer lines. Some of these lines may use shielded cable, but unless that cable is also well grounded, it may carry RF energy to sensitive devices with equally poor grounds.</p>
+  <p>Sensitive devices, including control inputs for the rig, do not require very much energy to show signs of interference. If all else fails, you can try the system shown in <b>Fig. 12</b>. At the building or shack entry, install a 1:1 choke of ferrite beads, following the designs of W2DU. The choke acts as a balun, converting the balanced line to the unbalanced coax. From the coax connector shell, run a very short earth-ground line. Ideally, the choke should go outdoors, but modern building construction may require immediate indoor installation at the entry point. Between the choke and a single-ended network tuner, run less than 20' of the largest, lowest-loss coax that you can obtain.</p>
+  <center>
+    <img src="abd-12.gif" width="375" height="225" border="2" vspace="5" hspace="9">
+  </center>
+  <p>The system shown will generally eliminate most unwanted RF energy transfers if the feeders have not already coupled into house wiring due to improper dress. It bypasses the 4:1 balun in the tuner, avoiding that loss source. However, the system has losses of its own. The 1:1 choke will show losses with high impedances having significant reactive components. The coax will also show some loss. However, if the length is 20', the line losses will usually be fairly small. For example, at 30 MHz, 20' of RG-213 will show a 1.1-dB loss with a 10:1 SWR. A shorter run, lower frequency, or lower SWR will result in lower coax losses. There are also cables with even lower losses. Do not use thin cables like RG-58 for this run, regardless of the operating power level. This system is not ideal, but simply a measure of last resort for very tough cases of RF interference. Before employing this or other radical systems, you should first use the earlier guidelines to optimize the feeder and tuner installation.</p>
+  <p>These notes do not answer every question that we can ask about the all-band doublet. However, I hope the 10 common questions that we have tackled give you a good start for reasoning out answers for yourself.</p>
+  <p><b>Notes</b></p>
+  <p>1. <b>Fig. 2</b> follows Stutzman and Thiele, <i>Antenna Theory and Design</i>, 2nd Ed. (Wiley & Sons, 1998), p.57, although the treatment of the dipole in very short or longer forms is similar in Kraus and in Balanis. (Kraus, especially, (<i>Antennas</i>, 2nd Ed.) is careful not to label longer center-fed wire antennas as dipoles).</p>
+  <p>2. Compare, for example, 2 editions of <i>The Radio Amateur's Handbook</i>: the 13th or 1936 edition, p. 278, and the 24th or 1947 edition, p. 207.</p>
+  <p>3. The inverted-L is a highly usable antenna for general-purpose communication, but requires an entirely different discussion. See "Straightening Out the Inverted-L" in the <i>2005 Proceedings of FDIM</i>, produced by QRP ARCI.</p>
+  <center>
+    <img src="../images/colorbar.gif" width="540" height="4">
+  </center>
+  <p><i>Updated 6-3-2005. &COPY; L. B. Cebik, W4RNL. Data may be used for personal purposes, but may not be reproduced for publication in print or any other medium without permission of the author.</i></p>
+  <center>
+    <img src="../images/colorbar.gif" width="540" height="4">
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