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<title>Antenna Options Index (19 Articles)</title>
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<h1><img src="../images/qex.gif" align="left" width="150" height="83" border="2" vspace="5" hspace="25"> <img src="../images/w4rnlwwr.gif" align="right" width="99" height="83" border="2" vspace="5" hspace="25"><i>Antenna Options</i></h1>
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<h3>L. B. Cebik, W4RNL</h3>
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<p>When asked to become a contributing editor for <i>QEX</i>, ARRL's technical experimenter's journal, I developed the "antenna options" theme as a foundation for my irregular contributions. Each episode in the series tries to lay out the options that an antenna builder, designer, or user has over some defined territory. Sometimes the options are purely practical--as practical as selecting element or boom materials. Sometimes the options match antenna designs to operating goals. Sometimes the options involve modeling software and its use. Occasionally, a topic requires more than one episode to finish (although we never really exhaust our options). However, the irregular publication schedule limits the number of columns devoted to a single topic. All items on the list below appear with permission, although they are copyrighted by ARRL. The material originally appeared in <i>QEX: Forum for Communications Experimenters</i> (<a href="https://www.arrl.org/qex" target="_blank">www.arrl.org/qex</a>). The episodes will appear at this site only after they have appeared in <i>QEX</i>.</p>
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<ul>
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<li><b><a href="ao1.html">No. 1: A Yagi Case Study: Part 1, Design Options</a></b></li>
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<li><b><a href="ao2.html">No. 2: A Yagi Case Study: Part 2, Element Material Options</a></b></li>
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<li><b><a href="ao3.html">No. 3: A Yagi Case Study: Part 3, Construction Options</a></b></li>
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<li><b><a href="ao4.html">No. 4: Antenna Modeling Software: Part 1</a></b></li>
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<li><b><a href="ao5.html">No. 5: Antenna Modeling Software: Part 2</a></b></li>
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<li><b><a href="ao6.html">No. 6: Do I Need More Gain?</a></b></li>
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<li><b><a href="ao7.html">No. 7: Wire and the HF Horizon: the Ys and Wherefores</a></b></li>
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<li><b><a href="ao8.html">No. 8: How Wide is Wide?</a></b></li>
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<li><b><a href="ao9.html">No. 9: A Broadside of Vertical Wires</a></b></li>
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<li><b><a href="ao10.html">No. 10: Horizontal Bi-Directional Wires</a></b></li>
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<li><b><a href="ao11.html">No. 11: From Two to One</a></b></li>
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<li><b><a href="ao12.html">No. 12: Narrowband NVIS Antennas</a></b></li>
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<li><b><a href="ao13.html">No. 13: NVIS Antennas for Special Needs</a></b></li>
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<li><b><a href="ao14.html">No. 14: Reversible Wire Beams for Lower HF Use</a></b></li>
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<li><b><a href="ao15.html">No. 15: Circularly Polarized Aimed Satellite Antennas</a></b></li>
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<li><b><a href="ao16.html">No. 16: Horizontally Polarized Omni-Directional Antennas: Some Compact Choices</a></b></li>
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<li><b><a href="ao16a.html">No. 16a: Horizontally Polarized Omni-Directional Antennas: Some Larger Choices</a></b></li>
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<li><b><a href="ao17.html">No. 17: Beam-Matching</a></b></li>
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<li><b><a href="ao18.html">No. 18: Reflections on Reflectors</a></b></li>
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</ul>
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<p>The versions of the episodes that appear here are in HTML format. Hence, they will diverge in appearance from the versions appearing in <i>QEX</i>. Occasionally, I shall have to change some wording to suit the arrangement and progression of graphics and tables.</p>
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<p><i>Updated 03-05-2008. © L. B. Cebik, W4RNL. Data may be used for personal purposes, but may not be reproduced for publication in print or any other medium without permission of the author and ARRL.</i></p>
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<p><a href="../radio.html"><img src="../images/return.gif" align="bottom" width="40" height="40"> Return to Amateur Radio Page</a></p>
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<p class="topics-footer">Topics: <a href='../topics.html#magazine-columns'>Magazine Columns</a></p><br>
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<title>AO 01: A Yagi Case Study: Part 1, Design Options</title>
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<h1 align="center"><img src="../images/qex.gif" align="left" width="150" height="83" border="2" vspace="5" hspace="25"> <img src="../images/w4rnlwwr.gif" align="right" width="99" height="83" border="2" vspace="5" hspace="25"><i>A Yagi Case Study<br>
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Part 1: Design Options</i></h1>
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<h3 align="center">L. B. Cebik, W4RNL</h3>
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<p>"This antenna is the best thing since sliced bread." Such is too often the claim made for antennas by individual builders and commercial makers alike. I'll bet those who make this and similar claims have not stopped to consider that for many meals, sliced bread is exactly the wrong bread to serve. I know a nice little restaurant that serves excellent soups in freshly baked bread bowls. I would not eat there if they tried to serve the soup in sliced bread.</p>
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<p>So, too, with antennas. For any general application, we have options. Only when we evaluate those options against our specific requirements and our situational limitations can we decide on the best antenna for the circumstances. Notice that the result is not simply "the best antenna." It is the best antenna for the given job and the conditions under which it will have to do that job.</p>
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<p>Since I cannot know every circumstance in which amateurs set up antennas, I cannot say what the best antenna is for any amateur activity. But I can use the space that QEX has allotted to me to discuss some options and alternatives for specific tasks. In small spaces, I cannot cover every possible option and certainly not all of the details that attach to each option. However, I can (hopefully) begin a thinking process that may ultimately let you make the best final decisions for yourself. The options that I have in mind are not brand-A vs. brand-B commercial offering. I do not have the appropriately rated test range for this kind of discussion. Instead, I shall look at options among antenna types, antenna construction, matching systems, etc. that one might face in deciding what to build for oneself.</p>
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<p>In virtually all areas of antennas, there are facets of design and performance that we easily overlook, and many of them have an important place in our decision-making processes. To make the process even more concrete, let's look at the myriad of options that attach to a seemingly simple case study that I call "A Tale of 3 Yagis."</p>
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<p>Yagi arrays require the antenna-builder to make three major decisions on the way from idea to reality.</p>
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<li>1. What design is best?</li>
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<li>2. What material is best for the elements?</li>
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<li>3. What assembly method is best?</li>
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<p>These three questions ultimately rest on another: what are the uses, purposes, or goals for the antenna? How all of these questions inter-relate is part of the motivation for this set of notes.</p>
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<p>To keep our work confined within a space that we can control, I shall examine only 2-meter Yagis. Within that space, I shall further assume that the user will take the antenna into the field for one or more of a variety of portable operations. To make matters even simpler, I shall restrict the discussion to 3-element Yagis with 30" boom lengths or less.</p>
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<p>Even with these restrictions, we still have design choices. We may select a narrow-band, high-gain design to maximize potential for point-to-point communications. Alternatively, we may select a design with a very high 180-degree front-to-back ratio for direction finding uses. Finally, we may select a design that covers the entire 2-meter band with an exceptionally low 50-Ohm SWR. For each design, there are trade-offs that we shall examine along the way.</p>
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<p>Once we select a design, we need to select the material for the elements. If we choose to use rod or tubing for the elements, we may simply optimize the design for the element diameter we wish to use. In fact, the design information provided in the tables will cover most common 2-meter rod and tubing sizes. However, my e-mail regularly poses questions about the use of non-standard materials, such as flat stock, L-stock, whips, and tape. Therefore, we shall spend a bit of time looking at a technique for determining what adjustments we might have to make for some of these materials. I shall provide some data that emerged from my own use of these techniques, but the techniques themselves will be your better guide to handling materials that I have not imagined.</p>
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<p>Finally, we shall look at a few methods of overall assembly that are suitable for the element material, the overall size of the antenna, and the intended use. For such a short boom, there is no reason not to use a non-conductive boom. In most cases, the choice will be between PVC and fiberglass, with PVC being easier to find and somewhat more versatile.</p>
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<p>In the end, these notes are just a sample of a thought process you can and should extend to other bands, other designs, and other operating purposes.</p>
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<h4>Part 1: The 3 Yagi Designs</h4>
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<p>The first step in our tale of 3 Yagis is to describe the Yagis themselves. There is a high-gain, narrow-band version, a maximum-front-to-back ratio version, and a very-wide-band version. A papa bear, mama bear, and chubby baby bear analogy in these characterizations is likely not accidental.</p>
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<p>For each antenna design, the tables will provide detailed dimensions for a variety of element diameters from 1/8" up to 1/2", in readily available rod and tube sizes. The material may be aluminum (recommended for its low weight and high strength), brass, and copper. The performance figures are based upon aluminum, although changing the material will not alter the performance in any detectable way.</p>
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<p>The tables also provide performance data from NEC-4 models at the design frequency. All designs attempt to achieve a minimum of 20-dB 180-degree front-to-back ratio across the listed passband. However, the particular design will reveal variations in where, within the operating passband, the maximum front-to-back ratio occurs. All 3-element Yagi designs show a gradual increase in gain across the operating passband.</p>
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<p align="center"><img src="ao1-1.gif" width="514" height="232" border="2" vspace="5" hspace="9"></p>
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<p><b>Fig. 1</b> shows the relative proportions of the 3 Yagi designs, using the 3/8"-diameter element versions as the basis for the sketch. The high-gain version has nearly equal spacing between elements and an almost uniform taper of the element outer tips. (The departure of the spacing and taper from uniformity is essential to achieving the performance.) The maximum 180-degree front-to-back ratio design preserves a similar driver-reflector structure, but shortens the length and spacing of the director to achieve the deep rear null. Both antennas have feedpoint impedance near 25 Ohms. The very wide-band version requires a 50-Ohm feedpoint and therefore widens the reflector-to-driver spacing. Note also the relatively short director.</p>
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<p>1. <i>High Gain: 144.5 MHz high gain 3-element Yagi</i>: The high-gain Yagi is designed for maximum gain with a reasonable boom length and modest bandwidth. It will cover about 2 MHz of the 2-meter band if the design frequency is moved from 144.5 MHz to 145 MHz. However, its present design recognizes that most point-to-point activity is in the first MHz of the band. So the design frequency is set at 144.5 MHz. Within the first MHz, the SWR is less than 1.2:1 for any listed element diameter. Although horizontal operation is the norm for the low part of 2 meters, the antenna is equally operable horizontally or vertically.</p>
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<p>We should note here that we can obtain higher gain from 3 elements with a longer boom and by setting aside the desired 20-dB front-to-back ratio. However, such designs usually also have much lower feedpoint impedances that may be difficult to match to a 50-Ohm cable without undue loss in the connections. The high-gain design used here is therefore a compromise among the many operating parameters involved in the directional antenna.</p>
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<p>The feedpoint impedance is resonant at about 25 Ohms. This arrangement is intentional to avoid the need for excessive numbers of mechanical connections at the antenna proper. A 1/4 wavelength section of RG-83 (35-Ohm) or a parallel section of RG-59 (or similar 70-Ohm) coax will provide a match to the 50-Ohm main cable. Cut the section for 144.5 MHz, allowing for the cable's specific velocity factor. Alternatively, one may modify the design to shorten the driver so that it shows about 25 Ohms of capacitive reactance. Then, a hairpin or gamma match becomes applicable.</p>
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<p>It is not possible to let the maximum front-to-back ratio of the high-gain, narrow-band design coincide with the design frequency for all element diameters. The smaller the element diameter, the more likely the maximum front-to-back ratio is to fall below the design frequency. However, the front-to-back ratio exceeds 22 dB from 144 to 145 MHz for all versions of the design.</p>
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<p align="center"><img src="ao1-t1.gif" width="599" height="203" border="2" vspace="5" hspace="9"></p>
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<p align="center"><img src="ao1-t2.gif" width="560" height="198" border="2" vspace="5" hspace="9"></p>
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<p><b>Table 1</b> provides dimensions for the design using round elements from 1/8" up to 1/2" in diameter. <b>Table 2</b> shows the modeled free-space performance at the design frequency--144.5 MHz--for each size material. Simplified figures for the edges of the 1-MHz operating passband appear in <b>Table 3</b>, while <b>Table 4</b> suggests the overall usable operating bandwidth with 25-Ohm SWR values at 144, 145, and 146 MHz. If the builder uses a 1/4 wavelength matching section for a 50-Ohm coaxial feedline, the 50-Ohm SWR at the junction of the matching section and the main feedline will be similar. <b>Fig. 2</b> shows free-space E-plane and H-plane patterns for the array at the design frequency. These patterns replicate the pattern shapes when the antenna is used over ground in the horizontal and vertical positions, respectively.</p>
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<p align="center"><img src="ao1-t3.gif" width="634" height="205" border="2" vspace="5" hspace="9"></p>
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<p align="center"><img src="ao1-t4.gif" width="349" height="162" border="2" vspace="5" hspace="9"></p>
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<p align="center"><img src="ao1-2.gif" width="400" height="225" border="2" vspace="5" hspace="9"></p>
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<p>The charts show clearly that the bandwidth for any particular characteristic tends to increase with an increase in element diameter. However, note that each increase in element diameter requires a change in element spacing as well as element length to sustain the performance curves over the 144-145-MHz passband. Most of the change in element spacing occurs with respect to the driver and reflector, since this spacing, relative to a given element diameter, largely determines the feedpoint impedance of the array, once we have set the performance values with the spacing and length of the director.</p>
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<p>To change the design frequency, scale both element lengths and element spacing. Take the ratio of the old (144.5 MHz) frequency to the new frequency and multiple all dimensions in the tables by the result. If the scaling is within the 2-meter band, no element-diameter adjustment is necessary. If the frequency ratio is greater than about 1.2:1 or less than 0.8:1, then element-diameter scaling is necessary to retain the performance characteristics.</p>
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<p>2. <i>Maximum Front-to-Back: 146 MHz maximum front-to-back 3-element Yagi</i>: The applications for a high-gain design are obvious. A maximum front-to-back ratio design has more limited application, for example in the field of amateur radio direction finding. The antenna should have sufficient gain to locate the desired signal and a sufficiently sharp and deep rear null to provide a reliable bearing toward the target transmitter. Although the horizontal pattern for any parasitic beam will include rear, quartering lobes, the vertical pattern will show a clear single null. Most direction-finding activities use vertical polarization.</p>
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<p>I have set the design frequency for the maximum front-to-back ratio at 146 MHz, because there is no absolute standard for direction-finding frequencies. However, for operation within the 2-meter band, one may scale the dimensions according to previously given principles without concern for scaling the element diameter. However, for scaling outside the limits of the 2-meter band, one should also scale the element diameter.</p>
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<p>Although not radically finicky, the maximum front-to-back ratio for a Yagi design is a narrow-band phenomenon. Hence, one should construct the antenna for the frequency of intended use. As the performance tables will show, the front-to-back ratio decreases steadily off-frequency until the design shows no distinct null. However, the design has sufficient gain and front-to-back ratio to make it a useful performer for other purposes.</p>
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<p>The feedpoint impedance for this design is set for about 25 - j 25 Ohms. The models all use an identical shorted transmission line stub across the feedpoint to simulate a hairpin match. Hence, the SWR curves are for 50 Ohms. The modeled feedpoint resistance is actually close to 27 Ohms, and the required inductive reactance of the hairpin is 54 Ohms. You may construct a U-shaped hairpin for the antenna by calculating the characteristic impedance for the spacing and wire diameter used. Then the length follows standard shorted-transmission line equations found in <i>The ARRL Antenna Book</i>, Chapter 24. A normally good construction method for the hairpin it to choose a distance between the parallel lines that is equal to the spacing between the driver terminals.</p>
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<p>As an example, AWG #14 wire (0.0641" diameter) has a 400-Ohm characteristic impedance at a center-to-center line spacing of 0.901". A shorted stub or hairpin made from this line would need to be 1.73" long to achieve 54 Ohms inductive reactance at 146 MHz. The final adjustment requires care, since the terminal structure at the feedpoint normally introduces some reactance that may add to or subtract from the amount provided by the hairpin. Lower characteristic impedances yield longer stubs for the same reactance. Narrowing the line spacing or fattening the conductor will lower the characteristic impedance. The goal is a hairpin that is short enough to be sturdy in field use but not so short as to make the final feedpoint adjustment too finicky.</p>
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<p>If you build this design for its intended purpose, general field adjustment also requires care. Contrary to most received wisdom, the reflector is not chief source of the front-to-back ratio in the design. The reflector is relatively insensitive and serves primarily to establish the feedpoint impedance by virtue of its length and spacing from the driver. The most sensitive element relative to establishing the ideal front-to-back ratio will be the director. The director length will be more sensitive than its spacing from the driver, although both dimensions deserve the label "sensitive."</p>
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<p align="center"><img src="ao1-t5.gif" width="600" height="204" border="2" vspace="5" hspace="9"> <b>Table 5</b> provides dimensions and <b>Table 6</b> supplies performance figures based on NEC-4 models for element sizes ranging from 1/8" to 1/2". In all of the designs in these notes, the dimension values presume a non-conductive boom or the use of mountings that insulate and isolate the elements from the influences of a conductive boom. See "Scaling and Adjusting VHF/UHF Yagis" at my web site for notes on adjusting element lengths for insulated through-boom construction (<a href="../vhf/scales.html">Scaling and Adjusting VHF/UHF Yagis</a>). However, for boom lengths under 30" or so, and for direct-feed drivers, there is little reason to use a conductive boom.</p>
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<p align="center"><img src="ao1-t6.gif" width="584" height="720" border="2" vspace="5" hspace="9"></p>
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<p>The performance figures show the increasing performance--although very gradually increasing--as we increase the element diameter and also adjust both element length and spacing to optimize performance for each new element size. This clear picture emerges largely because the design aligns the design feedpoint impedance and the maximum front-to-back ratio at the same frequency. The only performance figure for which variations make no difference is the 146-MHz front-to-back ratio: I ceased optimizing when this value exceeded 50 dB.</p>
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<p align="center"><img src="ao1-3.gif" width="400" height="225" border="2" vspace="5" hspace="9"></p>
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<p><b>Fig. 3</b> shows free-space E-plane and H-plane patterns, which replicate the patterns you will obtain when using the antenna in a horizontal or vertical orientation, respectively. Over ground, expect the vertical pattern to have less gain but a wider beamwidth than the horizontal pattern. However, when used vertically, the rear quadrants will show a single deep, sharp null.</p>
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<p>3. <i>Very Wide-Band: 146 MHz very wide-band 3-element Yagi</i>: The third design stresses smooth performance over the entire 2-meter band with a 50-Ohm SWR of less than 1.2:1. The design achieves this goal by using a variation of a W6SAI design from the 1980s, modified for 2 meters and for the full range of rod and tube diameters. Although the gain is about a full dB less than the high-gain model, the wide-band Yagi will provide roughly equal performance anywhere in the band. Thus, it is fit for flipping from horizontal to vertical and back again as we change to and from point-to-point and FM repeater operations.</p>
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<p>The Yagi uses a direct 50-Ohm feedpoint with no matching network required (although common-mode current suppression measures are advisable). The need to establish the SWR curve as the primary design goal has consequences as we increase the element diameter. The front-to-back ratio shows improvement with each larger element, but the low end of the band does not quite make the 20-dB level. Very quickly in the sequence, the gain ceases to increase with increasing element diameters. Although not clearly apparent in the performance figures, both the peak gain and the peak front-to-back ratio occur at ever higher frequencies. Above the smallest element size, the peak front-to-back ratio occurs above the upper end of 2 meters.</p>
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<p align="center"><img src="ao1-t7.gif" width="602" height="203" border="2" vspace="5" hspace="9"></p>
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<p align="center"><img src="ao1-t8.gif" width="584" height="720" border="2" vspace="5" hspace="9"></p>
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<p>Nevertheless, the very wide-band 3-element Yagi is a true general utility antenna for use anywhere in the band. <b>Table 7</b> provides the dimensions for elements ranging from 1/8" to 1/2" in diameter. <b>Table 8</b> supplies the modeled performance figures. <b>Fig. 4</b> gives us the shape of patterns when we use the antenna horizontally and vertically. All previous notes about scaling this antenna to other frequency bands are applicable with the very-wide-band Yagi.</p>
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<p align="center"><img src="ao1-4.gif" width="400" height="225" border="2" vspace="5" hspace="9"></p>
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<p>The very wide-band version of the 3-element Yagi completes the family of designs that we shall consider. Other variations may be possible, but these 3 cover the major performance parameters with which amateurs are most concerned: gain, front-to-back ratio, and operating or SWR bandwidth. One might further optimize the designs, but the level of optimizing used here gets the most out of each design that we can for each size element.</p>
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<p>The next step--assuming that one of these designs will meet an operating need--is deciding upon the element material. Our choices are not merely the material type, for example, copper, aluminum, or brass. They also include materials other than the uniform round tubes that our initial design models have used. We shall explore some of our options next time.</p>
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<p><i>Updated 01-12-2006. © L. B. Cebik, W4RNL. This item first appeared in</i> QEX<i>, Jul/Aug, 2004, pp. 55-59. Reproduced with permission. Copyright ARRL (2004), all rights reserved. This material originally appeared in</i> QEX: Forum for Communications Experimenters <i>(<a href="https://www.arrl.org/qex" target="_blank">www.arrl.org/qex</a>).</i></p>
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<p><a href="ao.html"><img src="../images/return.gif" align="bottom" width="40" height="40"> Return to series index page</a></p>
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<p class="topics-footer">Topics: <a href='../topics.html#vhf-uhf'>VHF / UHF</a>, <a href='../topics.html#yagi'>Yagi</a></p><br>
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<title>AO 10: Horizontal Bi-Directional Wires</title>
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<h1><img src="../images/qex.gif" align="left" width="150" height="83" border="2" vspace="5" hspace="25"> <img src="../images/w4rnlwwr.gif" align="right" width="99" height="83" border="2" vspace="5" hspace="25"><i>Horizontal Bi-Directional Wires</i></h1>
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<h3>L. B. Cebik, W4RNL</h3>
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<p>In the last set of notes, we discussed some of the options for bi-directional wire broadside arrays based on the vertical dipole. Since we should be fair to horizontal wires, this collection of notes will focus on bi-directional wire arrays that have their roots in the horizontal dipole. Just as the last collection was incomplete, so this one will be even further from exhaustive. Once we have some basic arrays available to use, the permutations and combinations form an endless progression. However, we can look at the various types of arrays.</p>
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<p>To be effective, a horizontal array should be well above ground. Although we might disagree on the minimum height for horizontal arrays, placing all arrays at 1 wavelength above ground for the top wire (if the array has a vertical dimension) will keep us well above the minimum. In the preceding notes, I set a height of 50' as a limit of sorts, since most serious array users can usually arrange 60' support posts, trees, or surplus telephone poles. The physical height restriction leads us to 15 meters, more specifically, a uniform test frequency of 21.225 MHz for all comparative horizontal array data. A wavelength at this frequency is 46.34'. To unify further the modeling conditions, the ground will be average, that is, have a conductivity of 0.005 s/m and a relative permittivity of 13. All antenna elements will use AWG #12 (0.0808" diameter) copper wire. As a result, we can more directly compare the reported performance characteristics for the antennas included in these notes.</p>
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<p>Traditionally, we categorize arrays as collinear, end-fire, or broadside. These categories are not mutually exclusive, but they do give us a convenient way of grouping our subject antennas. However, we must begin with a baseline.</p>
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<p><b>The Standard Antennas</b></p>
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<p>No antenna is more fundamental to amateur (and other) antenna work than the 1/2 wavelength resonant dipole. <b>Fig. 1</b> shows the outline of the antenna. Superimposed on the antenna is the distribution of the current magnitude along the wire's length. (Adding the current phase would create difficult viewing problems with some of our antennas.) Note that the dipole is electrically resonant, which requires a physical length that is 0.486 wavelength for #12 copper wire at the specified height above average ground. The resonant length of a dipole will vary with the height above ground, especially below about 1.25 wavelengths.</p>
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<p>Horizontal wire antennas tend to have an elevation angle of maximum radiation (or TO angle) that depends on the antenna's height more than any other factor. Hence, all of the basic antennas whose performance reports appear in <b>Table 1</b> have a TO angle of 14°. The dipole data in the table give numerical meaning to the plots on <b>Fig. 1</b>. Note the wide (79°) beamwidth for a standard dipole. A triangle of dipoles would suffice to cover the entire horizon. The dipole gain of 7.6 dBi becomes the baseline against which we may measure the gain of all subsequent arrays.</p>
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<p>The second standard antenna can go by several names. The table and <b>Fig. 2</b> both refer to a 1 wavelength center-fed doublet. An equally apt way to refer to the antenna is as a collinear array of 2 half wavelength wires. The current magnitude curve on the antenna outline shows why this name has good sense, since we observe two complete cycles of current rise and fall. Like the dipole, the physical length will be shorter than the electrical length. For the test conditions, we obtain resonance with a length of 0.956 wavelength of wire. Finding the resonant length of a center-fed 1 wavelength wire is tedious in modeling and virtually impossible with real wire. The resonant point occurs at a very high value of resistive impedance, where the reactance changes in a very short space from very inductive to very capacitive. Hence, there is an air of artificiality about the reactance number in the table, although the resistive part is a good indicator of what to expect at the feedpoint. One way to bring the impedance down to a value that is more amenable to our coaxial ways is to use a 1/4 wavelength section of 450-Ohm line.</p>
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<p>The pattern and the gain value are both very real and usable. The 1 wavelength wire provides about 1.6 dB additional gain relative to a dipole under equal conditions, but at a cost in the beamwidth. The longer wire has a beamwidth of about 51°. A triangle of 1 wavelength wires will not cover the horizon.</p>
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<p>The third standard wire antenna is the extended double Zepp (EDZ). We operate this antenna off resonance. We sometimes say that the antenna is non-resonant, but there is another use for this term. Many traveling wave and frequency-independent antennas show no cyclical appearance of resonant frequencies as we operate the antenna over a large frequency range. Many texts call these antennas non-resonant in contrast to antennas like a center-fed wire of fixed physical length. This antenna is resonant in the sense of showing cyclical re-appearance of zero reactance at the feedpoint as we sweep through a wide frequency range. So we may be content in calling the EDZ an off-resonance antenna. <b>Fig. 3</b> shows the current magnitude distribution for a physical length of 1.25 wavelength. The antenna qualifies as a collinear array.</p>
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<p>As we increase the length of a center-fed wire beyond 1 wavelength, we find that the gain increases until the length passes the 1.25 wavelength mark. However, the azimuth patterns and the tabular data show us other dimensions of what is occurring. As we move from about 1.2 wavelengths to 1.25 wavelengths, the resistive and capacitively reactive components of the feedpoint impedance decrease. (The gain changes only by about 0.1 dB.) Offsetting the reduction in the feedpoint impedance is the growth of the sidelobes that give an EDZ azimuth pattern its distinctive look. With a length of 1.2 wavelengths, the sidelobes are about 13 dB lower in gain than the main lobe. By a length of 1.25 wavelengths, the sidelobes have grown by 3 dB. At a total length of about 1.5 wavelengths, the side lobes would be as strong as or stronger than the main or broadside lobe. As well, the beamwidth continues to shrink as we lengthen the wire and increase the peak gain. Hence, the EDZ represents a sort of limit to the length of collinear arrays unless they employ techniques to correct phasing along the wire.</p>
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<p><b>Collinear Arrays</b></p>
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<p>One technique that allows us to obtain increased gain with straight wires is to introduce at critical points along the wire a phase-changing stub. Consider a 2 wavelength center-fed wire, such as the one shown in <b>Fig. 4</b>. We see 4 complete half wavelength current cycles. However, if we had just used a continuous length of wire. The phasing of the current at the outer half wavelength sections would have been 180° out of phase for producing a simple broadside pattern. Indeed, we would see for such a wire a 4-lobe pattern that forms a cloverleaf. By introducing 1/4 wavelength shorted transmission-line sections, we change the phase of the current at the beginning or inner end of the outer sections so that the radiation adds to the radiation of the center section. The result is a narrow main beam (the width is 26°) with over 1 dB gain over the EDZ. The tabular data for the collinear arrays appears in <b>Table 2</b>, for comparison with the patterns in the relevant figures.</p>
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<p>Like a 1 wavelength wire, which is what we find at the center of our collinear array that uses 1/2 wavelength sections, the feedpoint impedance is very high. As well, the azimuth pattern shows small EDZ-like sidelobes that are normal to very closely spaced 1/2 wavelength sections--in this case, the end sections relative to the longer inner section of the array.</p>
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<p>We need not limit ourselves to using 1/2 wavelength sections as the basis for a collinear array. As shown in <b>Fig. 5</b>, 5/8 wavelength section will work as well and provide additional gain. The collinear array shown uses such sections and has a total length of 2.78 wavelengths to achieve a peak gain of 13.5 dBi, about 1.5-dB higher than the comparable array using 1/2 wavelength sections. However, the beamwidth has shrunk to 16°. This array is for point-to-point communications.</p>
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<p>The array shows twice as many sidelobes because we have--in effect--a doubled EDZ structure. The inner section is slightly over 1.5 wavelengths, which seems long for an EDZ. However, if you examine the current magnitude curve on the outline sketch, you will see that it requires the added current rise not only at the feedpoint, but also at the location of the phasing lines. Because these regions are already reactive, the phasing lines can be shorter to effect the required phase reversal--in this instance, about 0.15 wavelength. The feedpoint impedance is normal for an EDZ-type antenna. Moreover, despite the high increase in array complexity, it still shows a TO angle of 14° that owes to its 1 wavelength height above ground.</p>
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<p>In the models for both of the collinear arrays, the phase lines are physical wire structures and not NEC TL constructs.1 The non-radiating TL lines are most accurate at high current regions of an antenna where the current changes very slowly over appreciable distances along the wire. They become less accurate in regions of the antenna in which the current changes rapidly over short distances. Since both collinear arrays show relatively low currents at the phase-line locations, physical lines are necessary for best accuracy in the models. The modeled line wires should be the same diameter as the antenna element wire, and the spacing should not be excessively close.</p>
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<p>The final sample of a collinear array takes an odd turn--upward. The bi-square array is a center-fed 2 wavelength wire with the ends turned upward to form a square that is 1/2 wavelength on a side. However, the wire ends do not meet, but leave a gap. <b>Fig. 6</b> shows the general outline and the current magnitude curves for each of the 4 wires. Because we have set a 1 wavelength limit for the top structure, the low point for the array is about 0.3 wavelength above ground. The net effect is to raise the TO angle of the array as a whole. A useful rule of thumb for any array consisting of multiple wires in the vertical plane is that the effective height of the array is about 2/3 the way up the array from the low point to the high point. Hence, the effective height of the bi-square in the illustration is about 0.75 wavelength.</p>
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<p>The bi-square has a beamwidth of about 63°, although its gain is about halfway between the values for the 1 wavelength and the EDZ arrays, both of which antennas exhibit narrower beamwidths. If you compare the elevation pattern for the bi-square with those for the other two relevant wires, you will discover that the bi-square lacks most of the upper structure or second elevation lobes that characterize the elevation patterns of the single wire antennas. The bi-square lends itself to single-support mounting, and you may place two such antennas at right angles on the support, switching to the one that yields best signal strength.</p>
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<p><b>End-Fire Arrays</b></p>
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<p>If a set of elements radiates in the plane of the elements with the lobes broadside to the element wires, then we have an end-fire array. Among basic arrays, we need only one sample: the W8JK "flat-top" array, developed by John Kraus, W8JK. The antenna has many incarnations using various element lengths and forms and using many spacing distances. In general, the longer the elements, for a given spacing, the higher will be the gain. Equally generally, the closer the spacing, the higher the gain will be. Hence, every W8JK is a compromise based on best achieving a particular set of goals for the antenna.</p>
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<p><b>Fig. 7</b> shows the outline of the 15-meter version that we shall examine. It uses a pair of 1 wavelength elements (or collinear 1/2 wavelength elements) spaced 1/2 wavelength apart. Between the elements, we find two phase lines of equal length meeting at a central point where we attach the main feedline. Note that the elements require a 180° phase difference, so one and only one of the phase lines receives a single half-twist. The modeled impedance data in <b>Table 3</b> rests on the use of 450-Ohm phase lines with a velocity factor of 0.9. Other line values will produce other impedances. The 15-meter patterns in <b>Fig. 7</b> are very clean with a bi-directional gain similar to that of a 2-element reflector-driver Yagi.</p>
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<p>The selection of the element length and spacing for this sample becomes apparent from the patterns in <b>Fig. 8</b>. As we raise the operating frequency, the spacing increases to lower the gain, but the element lengths increase to raise it. Lowering the operating frequency produces opposite effects. The result is an array that one might use over several bands with remarkably similar performance. The remaining gain changes are largely a function of the antenna's changing height when measured in wavelengths.</p>
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<p>The W8JK first emerged in 1937. For other configurations, see Kraus' <i>Antennas</i>, 2nd Ed., p. 458, as well as innumerable articles in <i>QST</i> over the years since the antenna first appeared. The array also lends itself to becoming part of more complex array systems.</p>
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<p><b>Broadside Arrays</b></p>
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<p>If the main radiation lobes of an antenna are at right angles to the plane of the elements, then we have a broadside array. Standard HF horizontally polarized arrays tend to require a vertical plane for the elements for broadside use, although NVIS operations might set the element in a horizontal plane for straight-up radiation. The most common simple 2-element broadside array for 15-meter and other upper HF bands is the Lazy-H, shown in <b>Fig. 9</b>. Like the W8JK, the elements are 1 wavelength long and spaced vertically 1/2 wavelength apart. However, we feed the elements in phase, so we do not twist either of the two phase-lines. (An alternative feeding system brings the main feedline to the bottom element. The line connecting the elements receives a half twist to produce a 360° phase change in the 180° phase line. The resulting feedpoint impedance is very high and requires a matching section.) <b>Table 4</b> provides the modeled data for the center-fed version of the array. The impedance reports rest on the use of 600-Ohm phase lines having a velocity factor of 1.0.</p>
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<p>With elements at 1/2 wavelength and 1 wavelength above ground, the lazy-H out-performs the W8JK on 15 meters, even though the effective height is only a bit over 0.8 wavelength. The increased gain is partly a function of suppressing the upper lobe structure that we see in the W8JK pattern as a result of the lazy-H's vertical stack of elements at a 1/2 wavelength spacing. Like the W8JK, we can use the lazy-H on other bands, as shown both in the tabular data and the patterns of <b>Fig. 10</b>. Even though the elements are not spaced for maximum vertical lobe suppression on other bands, the elevation patterns show good reduction in higher angle lobes compared to a flattop array. As we lower the operating frequency, the height of the lower wire in the lazy-H grows closer to the ground when measured as a fraction of a wavelength. Hence, the bi-directional pattern for 30 meters does not appear, since it would be weaker than a simple dipole at 46' above ground and have a higher TO angle. Nevertheless, the antenna is operable on that band. Unlike the W8JK, the lazy-H shows considerable variation in gain as we move from one band to the next.</p>
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<p>The pattern for 28.5 MHz calls for a brief note. Using elements that are 1 wavelength long for 15 meters produces a 10-meter pattern with very strong sidelobes, since the elements are over 1.3 wavelengths long on that band. Instead of beginning with a standard 15-meter design, we might have design for 10 meters a lazy-H that uses 1.25 wavelength elements with 5/8 wavelength spacing--about 44' and 22', respectively. The result would not change the 10-meter gain significantly, but it would yield an EDZ-type pattern with lower sidelobe content. As well, the antenna would operate over the same frequency range as our sample. This is the so-called expanded or extended lazy-H.</p>
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<p>The lazy-H is the foundation for many more complex arrays. In principle, we may combine the lazy-H and the W8JK into a 4-element array. The lazy-H also finds extensive use in commercial array construction for short-wave broadcasting. Consider a vertical bay of elements fed in phase, with as many element at 1/2 wavelength spacing as vertical supports will allow. Now place further columns of such arrays next to each other. Finally, place a large screen between 1/4 wavelength and 1/2 wavelength behind the rows and columns of antennas. In the late 1920s, we would have called the antenna a billboard array. If we reduce the element lengths to about 1/2 wavelength and use special wide-band forms of dipole elements, we end up with the modern dipole directional array. By staggering the feed current phase angle from one vertical bay to the next, we can actually slew the direction of the main lobe without ruining the overall characteristics of the antenna pattern. This brief account of modern dipole arrays overlooks a myriad of both electrical and mechanical engineering feats necessary to implement such an array, but it does show the fundamental place of the lazy-H among broadside arrays.</p>
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<p>Our final sample broadside array is a curtain, one that passed out of use long ago. Nevertheless, it serves well as a sample of an antenna that continues to attract amateur attention for its pattern properties. In <b>Fig. 11</b> we have the outline of a 3-section Sterba curtain, first reported in antenna literature in 1931. Basic to the design are the 1/2 wavelength by 1/2 wavelength inner sections, with end sections that are 1/4 wavelength wide by 1/2 wavelength high. We may use as many inner sections as we desire, but a centered feedpoint requires an odd number. The lines between sections are critical: they must be 1/2 wavelength long parallel transmission lines with a half-twist in each line to keep the top and bottom wires in phase with each other. The sample places the bottom wire 1/2 wavelength above ground to keep the array within our maximum height limits.</p>
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<p><b>Table 5</b> provides the modeled data to accompany the patterns in <b>Fig. 11</b>. Like the lazy-H, the elevation pattern shows good suppression of higher-angle lobes. The figure shows two azimuth patterns to accommodate the two most common feedpoints for Sterba curtains: the center of the lower element structure and the corner. The table includes only one data set, since there is no significant difference in the gain, the beamwidth, or the feedpoint impedance. A 600-Ohm feedline would handle either central or corner feeding very effectively. With central feeding, we have a bi-directional pattern that is exactly broadside to the array itself. However, if we feed that antenna at a corner, we bend the pattern by nearly 5° in the direction of the feedpoint. For such a large array (even though only 2 wavelengths or about 92' long), copper losses are sufficient to distort the pattern direction, although one might compensate through a tedious task of making slight dimension alterations.</p>
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<p>Like a lazy-H that uses a bottom feedpoint and a 1/2 wavelength half-twist phase-line, the Sterba curtain is a monoband antenna. On other frequencies, it becomes simply a semi-random collection of wire. As well, the Sterba requires a great quantity of wire. Consider once more the collinear array that used half wavelength sections. If we stack a pair of these antennas vertically, the total array length and height will be the same as for a Sterba curtain. However, we require only a single phase-line between the two to connect the feedpoints to the main feedline. The resulting array (with wires at 1/2 wavelength and 1 wavelength heights) will have a gain of nearly 15 dBi, but with a very great saving in wire.</p>
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<p><b>Another Interim Conclusion</b></p>
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<p>For the wire fans, horizontal bi-directional arrays offer considerable potential in the form of relatively simple, reliable, and inexpensive arrays. In the upper HF region, such antennas have a long-term home if we use care in the ones we select. The lazy-H and the W8JK may be the most flexible arrays in terms of potential multi-band use. However, we must consider the impedances at the tuner end of the main feedline. In some cases, a low antenna feedpoint impedance may create significant line losses, even in parallel lines. We may need to change dual phase-line lengths and impedances to arrive at the best compromise efficiency level on all bands. Monoband arrays with very high impedances will likely require matching sections to arrive at the lower impedances of common coaxial or parallel transmission lines. Simple antennas can present not-so-simple feeding challenges.</p>
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<p>Likewise, we must use care in our selection of an array in terms of the gain-vs.-beamwidth question. If we had just one friend with whom we wish to communicate (or two friends at 180° separation), then we might use gain alone as our desideratum. However, in most cases, we shall want to balance gain against coverage. Hence, it is likely that we shall have to trade some gain to obtain adequate beamwidth.</p>
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<p>Just when we might think we have reached an ending, someone has whispered to me, "How do we convert a bi-directional array into a directional array?" It turns out that, even here, we have options.</p>
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<p><a href="ao10-models.zip">ao10-models.zip</a>.</p>
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<p><i>Updated 08-27-2006. © L. B. Cebik, W4RNL. This item first appeared in</i> QEX<i>, Sep/Oct, 2006, pp. 55-61. Reproduced with permission. Copyright ARRL (2006), all rights reserved. This material originally appeared in</i> QEX: Forum for Communications Experimenters <i>(<a href="https://www.arrl.org/qex" target="_blank">www.arrl.org/qex</a>).</i></p>
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<p><a href="ao.html"><img src="../images/return.gif" align="bottom" width="40" height="40"> Return to series index page</a></p>
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<p class="topics-footer">Topics: <a href='../topics.html#bruce-array-sterba-curtain'>Bruce Array / Sterba Curtain</a>, <a href='../topics.html#collinear'>Collinear</a>, <a href='../topics.html#dipole'>Dipole</a>, <a href='../topics.html#doublet'>Doublet</a>, <a href='../topics.html#edz-zepp'>EDZ / Zepp</a>, <a href='../topics.html#hf-upper'>HF Upper</a>, <a href='../topics.html#lazy-h'>Lazy H</a>, <a href='../topics.html#phased-array'>Phased Array</a>, <a href='../topics.html#w8jk'>W8JK</a></p><br>
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<h1><img src="../images/qex.gif" align="left" width="150" height="83" border="2" vspace="5" hspace="25"> <img src="../images/w4rnlwwr.gif" align="right" width="99" height="83" border="2" vspace="5" hspace="25"><i>From Two to One</i></h1>
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<h3>L. B. Cebik, W4RNL</h3>
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<p>In the preceding two episodes, we examined clusters of options for vertical and horizontal bi-directional wire arrays based on the dipole as the root antenna. Each cluster only sampled the options available, and we might easily add to the number of clusters. However, our goal was to present each cluster in an internally consistent environment to ease some of the difficulty in decision-making. Within each group, the individual can now use consistent performance data in combination with dimensional data, installation challenges, and budgetary cautions when deciding on the next antenna to build.</p>
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<p>Very early on, a perennial question arose: how does one convert a bi-directional array into a directional beam? Once more, we have options. Most handbooks treat the options in different chapters, so we rarely receive any kind of overview of available techniques. Although our space is small and therefore our samples will be few, we can partly correct the oversight.</p>
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<p>Consider two elements of similar size, at least one of which would serve as a bi-directional antenna. To create a directional end-fire beam from these two elements, we have to provide each element with the correct relative current magnitude and phase angle to yield the strongest possible forward lobe and the weakest possible rearward lobe. Similar considerations apply to both multi-element arrays and to large collinear arrays. As a convenience in sorting out how we accomplish the goal, we can separate the techniques into 4 general groups: parasitic methods, phasing methods, screen reflection, and traveling-wave termination. Not all techniques are equally apt to all antenna types and situations, but all have a place in the option set.</p>
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<p><b>Parasitic Methods</b></p>
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<p>Many hams associate parasitic elements almost solely with Yagi-Uda arrays based on the half wavelength antenna element. However, designers have long realized that parasitic elements will work with almost any bi-directional antenna. A parasitic element approximates the conditions that we may achieve directly by complex phasing systems, but it does its work solely by virtue of its length, diameter, and spacing from the original element. For a given spacing and diameter, if we shorten an element relative to the original one with a feedpoint (now called the driver), the new element will show a negative relative phase angle for its current and the main forward lobe will form in its direction. It has become a director. If we lengthen the element relative to the driver, then its current will show a positive phase angle relative to the current on the driver and the main forward lobe will be in the direction of the driver. The new element has become--by convention--a reflector. Reflector-driver arrays tend to show a wider operating bandwidth but poorer gain and front-to-back characteristics than optimized driver-director arrays. Hence, reflector-driver arrays are easier to tame, that is, easier to field-adjust into acceptable performance. We may fairly use these basic principles of parasitic element operation to create a large variety of directional beams.</p>
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<img src="ao11-1.gif" width="600" height="450" border="2" vspace="5" hspace="9">
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<p>Consider the half-square array, essentially a broadside array consisting of 2 vertical dipoles in an easily reproduced wire antenna. See <b>Figure 1</b>. The sketch shows the single half square along with 2 parasitic variations. The dimensions differ slightly from those used in our first array episode to yield a near-50-Ohm feedpoint impedance. However, we are still using AWG #12 copper wire, average ground, and a maximum height of 50'. The left-most plot shows the azimuth pattern with its slight misalignment due to the corner feedpoint. <b>Table 1</b> supplies the modeled data.</p>
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<img src="ao11-t1.gif" width="624" height="240" border="2" vspace="5" hspace="9">
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<p>We may add to the original array a reflector and optimize the vertical lengths for maximum front-to-back ratio with the selected spacing. The forward gain increases by about 3.2 dB with only a small narrowing of the beamwidth. The 180° front-to-back ratio is excellent. The rearward quartering lobes show a very slight imbalance, but not to a point of creating problems in using the array. The structure is very simple to replicate, and the feedpoint impedance remains "coax-ready."</p>
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<p>Rather than lengthen the reflector physically, we may also lengthen it electrically. One convenient method is to use a shorted length of transmission line to add to a driver-length element enough inductive reactance to create an effective reflector element. The sample uses 50-Ohm line at an electrical length of 17'. Suppose now that we add such a length (adjusted physically for the velocity factor of the selected coax) to the driver and bring it to a central point between the elements. We can install a remotely operated switch that converts one side into just an extension of the feed cable and the other side into the required shorted-stub. (Use a DPDT switch or relay to isolate the braids as well as the center conductors.) We now have a reversible half-square beam with virtually the same gain as the fixed version and a very useful rearward lobe set. The need to place the load and the feedpoint on the same side of the array creates a bit more imbalance in the pattern, but nothing fatal to the array's use.</p>
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<img src="ao11-2.gif" width="600" height="420" border="2" vspace="5" hspace="9">
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<p>We can also apply parasitic methods to collinear arrays, for example, the extended double Zepp (EDZ) shown in <b>Figure 2</b>. The sample EDZ uses a length just above 1.2 wavelength to reduce the sidelobes. Like our earlier horizontal bi-directional arrays, the sample uses AWG #12 copper wire and is 1 wavelength above average ground. <b>Table 2</b> provides the modeling data for both the single EDZ and two versions of a parasitic beam.</p>
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<img src="ao11-t2.gif" width="642" height="387" border="2" vspace="5" hspace="9">
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<p>The first version uses a full-size reflector element. Since the element will be capacitively reactive, we can incorporate electrical lengthening into the center loading inductive reactance and tune the element for maximum front-to-back ratio. We also have the option of adding a coil to the center of the driven element to achieve resonance and simplify matching. For example, with the same parasitic array set for resonance, we might use a 1/4 wavelength section of 75-Ohm cable as a matching section for a 50-Ohm main feedline. Providing the driver with inductive reactance at a normal Q range of 200 to 250 has its costs in terms of a slightly lower forward gain. However, the forward gain shows well over a 3-dB increase relative to the single EDZ element.</p>
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<p>We may also simplify the structure somewhat by using a split reflector element, with a section for each outer half wavelength of the driver. The front-to-back ratio will not be as high as with a full-size element properly loaded, but we save the trouble of having to field adjust a reflector inductor for maximum front-to-back ratio.</p>
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<p><b>Figure 3</b> shows two ways of obtaining reversible EDZ beams from the same reflector options. In both cases, we find decreases in the reported data from the model in one or another performance category, as shown in the bottom lines of the table. Mutual coupling between the active and inert drivers is stronger than in the case of comparable reversible-direction wire Yagis. Nevertheless, some combination of elements may satisfy a given set of operating needs.</p>
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<p><b>Phasing Methods</b></p>
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<p>We often employ phasing methods to improve the rearward null performance of an array. Rarely do phasing methods show very marked gain increases over parasitic methods applied to essentially the same array. In some cases, such as the fixed half-square beam, resorting to phasing techniques would only add considerable complication for little, if any, detectable improvement. In other cases, however, phasing treatment may show marked improvement.</p>
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<p>Phasing techniques essentially add an energy source to what we may achieve with parasitic methods. Parasitic techniques achieve the goal of setting the relative current magnitudes and phase angles to create a beam by virtue of mutual coupling between the elements. A driver-reflector Yagi, for example, rarely surpasses 12-dB front-to-back ratio. If we phase feed the two elements, we can create current magnitude and phase relationships that can easily raise the front-to-back ratio to 20 dB. Indeed, if we are willing to sacrifice some gain, we can create 180° front-to-back values as high as 50 dB--but only for the narrowest of bandwidths.</p>
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<img src="ao11-4.gif" width="550" height="220" border="2" vspace="5" hspace="9">
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<p>Many hams have a restricted view of phasing techniques. ON4UN's <i>Low-Band DXing</i> has perhaps the largest collection of practical phasing techniques. We may usefully but incompletely categorize the techniques into 3 groups, as shown in <b>Figure 4</b>. On the left are methods that use transmission-lines to achieve the requisite splitting of current and transforming the current magnitudes and phase angles to optimal values. The ZL-Special technique of using a single 1/8 wavelength line with a half-twist misled many builders into believing that line impedance transformation was the critical factor. However, the rate of current transformation along a line may not coincide with the impedance transformation. Particular set-ups of elements may require a forward, a rear, or an intermediate feedpoint. One part of the line may require a half-twist or not. Indeed, there are many situations in which available transmission lines or those we might construct simply will not provide the required current magnitude ratio and the phase angle difference that yields an effective beam.</p>
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<p>The most direct route around this problem is to employ phase-changing networks at each element, as suggested by the middle sketch. Due to the weight of such networks, we ordinarily employ them with ground-mounted vertical monopole systems. An alternative is shown on the far right. We may employ ground-mounted networks and use measured transmission-line lengths to each element. Current undergoes a full cycle in 360° of line (not in 180°, as is the case for impedance). However, the lines may have different lengths to add additional current and phase angle transformations to the output of the network or networks.</p>
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<img src="ao11-5.gif" width="600" height="340" border="2" vspace="5" hspace="9">
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<p>Network calculations require more space than we have for this overview, but we can demonstrate the potential benefits of a transmission-line-based phasing system with a single example. Consider 2 vertical dipoles set up for parasitic operation, as shown in <b>Figure 5</b>. The element center points are at equal heights, and the lowest approach to average ground for this 7.15-MHz array is 15'. With the 1" diameter aluminum elements used for the array, we obtain the patterns shown in the figure and the modeled performance figures that appear in <b>Table 3</b>. Like virtually all good 2-element driver-reflector parasitic systems, we obtain about 3.2-dB gain over a single vertical dipole. As we might expect from such an array, the maximum front-to-back ratio is just over 12 dB. With the element dimensions shown, the feedpoint impedance is close to 50 Ohms.</p>
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<img src="ao11-t3.gif" width="610" height="203" border="2" vspace="5" hspace="9">
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<p>Without changing any of the array dimensions, we may improve performance, especially with respect to the front-to-back ratio, as shown on the right side of <b>Figure 5</b>. The required phase line has two sections. The short section goes to the forward element. The longer section goes to the rear element and uses a half twist. The phase line sections consist of a foam dielectric coaxial cable with a 50-Ohm impedance and a velocity factor of 0.78. (A 0.66 velocity factor would yield physical lengths too short to handle the spacing between elements.) The net feedpoint impedance is about 30 + j13 Ohms, hence the need for a section of 35-Ohm cable (or parallel sections of 70-Ohm cable) to yield an impedance close to 50 Ohms. The specified length entails a velocity factor of 0.66. For the effort of figuring or finding a usable phasing system, we obtain a very small increase in gain and a very large improvement in the front-to-back ratio. At the same time, we maintain a coax-ready array.</p>
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<p><b>Planar or Screen Reflector Techniques</b></p>
|
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<p>Screen or planar reflectors date back to the 1920s and the billboard array. Once considered ungainly at HF, they find extensive use in the UHF region. The reflector is largely untuned, although extensive modeling has shown that gain reaches maximum when the limits of the screen at about 0.5 wavelength beyond the limits of the driving array in all dimensions. The screen is unlike the parasitic reflector, which does not reflect but instead has a size to yield the correct current magnitude and phase angle to yield a maximized forward and a minimized rearward lobe. The screen is a reflecting surface that acts largely (but not exclusively) according to principles derived from optics.</p>
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<img src="ao11-6.gif" width="705" height="400" border="2" vspace="5" hspace="9">
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<p>In the HF region, the short-wave broadcast industry makes extensive use of screen reflector arrays with banks of dipoles fed in phase for a main beam that is broadside to both the screen and the driving array. <b>Figure 6</b> shows the outlines of a very large array consisting of 3 rows and 3 columns of dipoles. Only a few of the support cables appear in the sketch. To have shown them all would have obscured the essential electrical elements of the array. It is possible to construct far smaller screen arrays to good effect. For example, a lazy-H is a bi-directional array that corresponds to a 1V-2H scheme in terms of the designations in <b>Figure 6</b>. Not only would a screen that is 2 wavelengths high by 2 wavelengths wide yield a directional signal, it would produce about 5 dB gain over the maximum gain of the basic lazy-H.</p>
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<img src="ao11-t4.gif" width="629" height="394" border="2" vspace="5" hspace="9">
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<p>The models used to test the capabilities of the dipole array with a screen reflector use commercial standards. Commercial dipole arrays strive to be broad-banded. The wire is AWG #10 copper (although many arrays use aluminum-coated steel). A combination of folded dipoles and a spacing of 0.3 wavelength yields a wider operating bandwidth, although it produces less than absolutely maximum gain. <b>Table 4</b> shows the modeled free-space performance of screened arrays from 1V-1H up to 3V-3H. Corresponding to the data entries are the azimuth patterns that appear in <b>Figure 7</b>. Each pattern rests on the use of a screen with dimensions optimized for the driving array size. (Commercial HF dipole arrays may skimp a bit on the screen size, and because the elements are horizontally polarized, the screen may use only horizontal wires. When a screen uses only horizontal wires, some reflector elements may exhibit parasitic properties, including high-than-normal current magnitude excursions.)</p>
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<p>The data and patterns are notable in several different ways. As we add elements vertically, the beamwidth does not change, although for each column, we find about the same gain increase with each added vertical bay. When we read the table and the patterns horizontally, we discover that each horizontal bay that we add reduces the beamwidth and increases the front-to-back ratio. One major advantage of the dipole array is that it allows the operator to select the best compromise between gain and beamwidth just by opting for an array size. In addition, by phasing each vertical bay in equal increments (for example, 30°, 60°, 90° for a 3-bay array), it is possible to slew the signal angle with respect to the plane of the array without major distortion to the pattern shape.</p>
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<p>In amateur use, a planar or screen reflector accepts a wide variety of drivers. The drivers may be either vertically or horizontally polarized and range from simple dipoles to more complex bi-directional arrays. Half-squares, rectangles, and side-fed quads are all applicable to a planar reflector to increase the forward gain and to yield a very good front-to-back ratio. In addition, for these single-feedpoint drivers, we can often find a spacing between the reflector and the driver that yields a desired feedpoint impedance, such as 50 Ohms, without sacrificing significant gain. One caution to observe is not to bother using an end-fire array in front of the screen. The screen will dominate so that the phased elements will not provide any significant further gain.</p>
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<p><b>Traveling-Wave Termination</b></p>
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||||
<p>Long-wire technology arose in the late 1920s and more especially in the 1930s in answer to the need for very precise trans-oceanic point-to-point communications. Edmond Bruce stands at the head of the class of antenna engineers for his invention of the "king" of long-wire antennas. Although these collinear arrays have largely passed out of service as installations strive to save real estate, amateurs still dram of wire antenna farms. Indeed, one humorist from past decades once specified his ideal DX antenna as a very large rhombic installed on a rotatable island. The basic long-wire shapes are the single long wire, the V, and the diamond or rhombic. Each of these shapes is essentially bi-directional. We obtain a directional beam by adding a terminating impedance at the array end opposite the feedpoint. In doing so we convert a standing-wave antenna into a traveling-wave antenna, although the conversion is imperfect.</p>
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<img src="ao11-8.gif" width="400" height="300" border="2" vspace="5" hspace="9">
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<p>We measure the basic single long-wire in wavelengths. The sample shown in <b>Figure 8</b> is 4 wavelengths long. As the patterns show, the long-wire has a split lobe. As we make the antenna longer, the lobes in each direction grow closer to the axis of the wire itself, but they never meet for almost all practical lengths. The sample long-wire antennas in these notes use 0.16" diameter (AWG #6) wire and are 1 wavelength above average ground at a test frequency of 3.5 MHz. The end-fed single wire has a lobe count that is twice the number we would expect of a center-fed wire of the same length. As the data in <b>Table 5</b> show, an unterminated or standing-wave long-wire has a natural imbalance of nearly 2 dB away from the feedpoint.</p>
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<p>When we add a terminating impedance (in this case 800 Ohms), we obtain the directional pattern shown in <b>Figure 8</b>. The gain drops significantly, not only because of the terminating resistor, but as well because each end of the array must have a wire to a common ground. The two vertical lines tend partially to fill the pattern nulls and thereby prevent the array from obtaining full gain. Nevertheless, we also obtain a very wide operating bandwidth. In fact, the SWR bandwidth is significantly wider than the practical bandwidth for the pattern shape.</p>
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<img src="ao11-9.gif" width="400" height="450" border="2" vspace="5" hspace="9">
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<p>The V array in <b>Figure 9</b> rests on the long-wire. The angle between the two legs is a function of the lobe angle of each leg--as shown by the lobes of the single long wire. Two lobes coincide down the centerline between lobes, resulting in a bi-directional pattern that is close to equal strength in both directions. The feedpoint is in series with the legs at their junction. If we wish to add a termination, we must run lines to ground, where we place separate terminating impedances, or we must run a line between the far ends of the V with the termination centered. In either case, we discover a decrease in gain relative to the unterminated array. However, the pattern is very strongly directional, and the SWR bandwidth far outstrips the practical operating bandwidth. With a V array, changing operating frequency changes the leg length as measured in wavelengths. Hence, the angle for an acceptable pattern has a limited frequency range of about 2:1.</p>
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<img src="ao11-10.gif" width="400" height="380" border="2" vspace="5" hspace="9">
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<p>With the possible exception of VK3ATN's moon bounce 2-meter rhombic in the 1960s, I am unaware that anyone has ever bothered to use an unterminated rhombic, as shown in <b>Figure 10</b>. However, I have included data on it in <b>Table 5</b> to compare with the far more common terminated version. A rhombic with 4 wavelength legs is, of course, twice as long overall as a V with 4 wavelength legs, but the width is about the same, since the same angle-based construction is involved. The terminating impedance of a rhombic (850 Ohms in this sample) is in series with the collinear array wires. Therefore, we see far less difference between the gain of the unterminated and the terminated versions. However, we can achieve very high front-to-back ratios.</p>
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<p>For commercial service, the major failing of all long-wire technology was the high level of the sidelobes, clearly evident on all of the patterns. The correct V or rhombic angle might combine two or more long-wire lobes, but it did little to suppress the other lobes in the long-wire pattern. For amateur serve, optimized V-beams and rhombics have two major drawbacks. Acreage is one of them. The other is the extremely narrow beamwidth. Nevertheless, the technique of using a termination to convert a bi-directional array into a directional beam deserves its place among the options available to us.</p>
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<p><b>Conclusion</b></p>
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<p>These notes have taken a somewhat different approach to converting a bi-directional array into a directional beam by providing an overview of the major techniques available to us. Not every technique is applicable to every bi-directional array that we might wish to convert into a directional beam. As well, in some cases, a technique might be applicable in principle but highly impractical to implement. In most cases, however, we likely will find more than one potential method of conversion and can weigh each against the other factors (for example, complexity, finickiness, budget, etc.) that go into our final building decisions.</p>
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<p><a href="ao11-models.zip">ao11-models.zip</a>.</p>
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<p><i>Updated 11-01-2006. © L. B. Cebik, W4RNL. This item first appeared in</i> QEX<i>, Nov/Dec, 2006, pp. 51-59. Reproduced with permission. Copyright ARRL (2006), all rights reserved. This material originally appeared in</i> QEX: Forum for Communications Experimenters <i>(<a target="_blank">www.arrl.org/qex</a>).</i></p>
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<p><a href="ao.html"><img src="../images/return.gif" align="bottom" width="40" height="40"> Return to series index page</a></p>
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<p class="topics-footer">Topics: <a href='../topics.html#broadcast'>Broadcast</a>, <a href='../topics.html#dipole-curtain-array'>Dipole Curtain Array</a>, <a href='../topics.html#edz-zepp'>EDZ / Zepp</a>, <a href='../topics.html#hf-lower'>HF Lower</a>, <a href='../topics.html#half-square'>Half Square</a>, <a href='../topics.html#phased-array'>Phased Array</a>, <a href='../topics.html#rhombic'>Rhombic</a>, <a href='../topics.html#vertical'>Vertical</a></p><br>
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<title>AO 12: Narrowband NVIS Antennas</title>
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<h1><img src="../images/qex.gif" align="left" width="150" height="83" border="2" vspace="5" hspace="25"> <img src="../images/w4rnlwwr.gif" align="right" width="99" height="83" border="2" vspace="5" hspace="25"><i>Narrowband NVIS Antennas</i></h1>
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<h3>L. B. Cebik, W4RNL</h3>
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<p>Dean Straw, N6BV, wrote an article for <i>QST</i> in December, 2005: "What's the Deal About 'NVIS'?" The article provides some excellent guidance for obtaining the best results from Near Vertical Incidence Skywave (NVIS) operation. The discussion limits itself to using a simple inverted-V antenna, which prompted the following notes. We have a number of options for potentially effective NVIS antennas. In this episode, we shall look at antennas that are narrow band, that is, antennas that cover one or part of one amateur band. We have enough to learn about them to occupy us fully.</p>
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<p><b>Figure 1</b> sketches the NVIS situation in the most general terms. Regular amateur operations seek to elevate antennas to provide low-angle radiation. Ionospheric refraction results in a skip zone, an area between the central station and the nearest communications target. In addition, many central stations have obstructions that limit the range of point-to-point communications methods. In both cases, directing a lower HF signal upward can result in a sufficient return to provide short to intermediate range communications. Many government services consider the NVIS frequency range to extend from 2 to about 10 MHz. As Straw notes, the 7-MHz region is most suitable for nighttime work, while the 80/75-meter band provides the best results for daytime operation by radio amateurs.</p>
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<p>Although you may set up many antennas for somewhat directional patterns, most operators strive to have an omni-directional antenna. Unfortunately, pure omni-directionality is hard to obtain with simple antennas. However, you can approximate a circular azimuth pattern by choosing the right antenna, as shown on the left in <b>Figure 2</b>. The elongated azimuth pattern shown on the right may also be useful. For simple wire antennas, the broader pattern is off the ends of the wire. If the installation area permits, you can go some distance in planning your coverage. The figure also shows a convention that I shall use in these notes: listing the broadside and the endwise half-power beamwidth. The closer these numbers are to each other, the more circular will be the pattern. The greater the difference, the more elongated that the oval pattern becomes. The ratio of one to the other is a useful measure of pattern circularity.</p>
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||||
<p>The final preliminary note concerns the antenna environment. We shall be looking at narrow-band antennas for use at the central station. In amateur terms, that generally means a durable home installation for which one may plan and then construct with care. The short-masted AS-2259 antenna is designed for field use, which might be a central station on a military battlefield. The antenna is useful to amateurs in Field Day and similar exercises. However, for long-term NVIS antennas, we can do far better.</p>
|
||||
<p><b>The Lowly Dipole</b></p>
|
||||
<p>The standard AWG #14 copper wire dipole will be our starting point for these NEC-4 modeling tests. All antennas will use average ground (conductivity = 0.005 S/m, relative permittivity = 13). The test frequency will be 3.9 MHz. A wavelength is about 252.2' at this frequency. The tabular data will be in fractions of a wavelength, so this number is handy for translating the information into numbers for physical planning. The trends that we uncover will be applicable throughout the lower HF range. <b>Figure 3</b> sketches the 4-dipole configurations that we shall examine.</p>
|
||||
<center>
|
||||
<img src="ao12-3.gif" width="550" height="350" border="2" vspace="5" hspace="9">
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||||
</center>
|
||||
<p>The first case of a dipole over bare ground has two goals. One aim is to see at what antenna height we obtain maximum upward gain. The second purpose is to put to rest a certain persistent myth about NVIS dipoles, namely, that a super-low height provides a gain advantage. <b>Table 1</b> provides expanded information on the performance of a dipole over bare ground at heights ranging from 0.05 wavelength (about 12.5' at 3.9 MHz) up to a quarter wavelength (63'). The table provides gain values from NEC-4 using the Sommerfeld-Norton ground calculation system (referred to as "high accuracy" in EZNEC). It also provides gain numbers reported by the only modeling program readily available during the early days of NVIS antenna analysis in the late 1980s and very early 1990s. That program was MININEC. As early as February, 1991, Roy Lewallen, W7EL, provided warnings to QST readers about the limitations of the MININEC simplified ground calculations system in his article "MININEC: The Other Edge of the Sword." Unfortunately, even today, many beginning modelers do not heed the warning. As the table shows, when we place a horizontal antenna below about 0.2 wavelength, MININEC reports an ever-inflating gain value. At the lowest height used in the table, the actual gain is 8 dB lower than the MININEC report. See the Straw article for the safety concerns and the supposed noise advantage of very low antennas.</p>
|
||||
<center>
|
||||
<img src="ao12-t1.gif" width="637" height="313" border="2" vspace="5" hspace="9">
|
||||
</center>
|
||||
<p>The table shows a gain peak with the antenna about 0.175 wavelength above ground. Although this height will be consistently the peak gain height for all of our simple antennas, heights from about 0.125 wavelength up to about 0.225 wavelength are perfectly acceptable. As we raise the antenna in small increments, we notice a slow rise in the endwise beamwidth, but a more rapid rise in the broadside beamwidth. Hence, we can go some way toward tailoring the circularity or elongation of the pattern simply by varying the height without seriously subtracting from the available gain.</p>
|
||||
<p>The remaining 3 configurations for a NVIS dipole reflect methods that some operators use or should use to improve performance. <b>Table 2</b> supplies the corresponding modeling data, but restricts the height range to values from 0.125 wavelength to 0.225 wavelength. The first supplemented dipole uses a single 1/2 wavelength wire at ground level below the dipole. This wire and all other antenna supplements use modeled heights of 0.001 wavelength above ground so that the models will run on both NEC-2 and NEC-4. The installer's goal is to create a virtual Yagi pointed upward. Contrary to expectations, the table shows a very limited improvement in maximum gain, with the best improvement at the lowest height. Ground reflections do not occur just below the antenna wire, but over a very wide area in all directions from the antenna.</p>
|
||||
<center>
|
||||
<img src="ao12-t2.gif" width="640" height="640" border="2" vspace="5" hspace="9">
|
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</center>
|
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<p>Studies of HF dipole arrays used for short-wave broadcasting and of VHF/UHF planar reflector arrays show that a conductive screen forms a very useful reflecting surface based on principles derived from optics. Such screens perform best when they extend at least 1/2 wavelength beyond the driven elements in all directions. The model with the screen in <b>Figure 3</b> and in <b>Table 2</b> uses a screen that is 1 wavelength by 1 wavelength on a side. The cells are 0.1 wavelength on a side. To simulate a solid screen, the wire would have to be very thick and using that wire would prevent the screen from sitting 0.001 wavelength above ground. So I reduced the wire size to a 1" diameter. The reduced wire size leaves the screen "holy" and reduces the reported gain. However, it may also better reflect the likely amateur use of inexpensive materials like chicken wire in which the junctions are not durably connected. The tabulated data shows a nearly constant improvement over the dipole and single-wire reflector. It also shows a decreasing improvement over the dipole above bare average ground. Still, the peak gain height remains at 0.175 wavelength. The broadside beamwidth shows a 4°-5° reduction with the screen in place, but the endwise beamwidth does not change at all.</p>
|
||||
<p>We can simulate a full screen with a series of wires at ground level if we use enough of them. The final configuration uses 9 AWG #14 copper wires at a height of 0.001 wavelength. The wire spacing is 0.1 wavelength. Each wire is only slightly longer than the dipole itself. With the 9 wires forming a field that is 0.8 wavelength long, the final section of <b>Table 2</b> shows performance virtually identical to the performance with a full screen. Smaller numbers of wires or total field sizes produce lesser performance levels. (Since the single reflector proved so ineffective and since the screen is simpler to model than the 9-wires, analyses of other simple antenna will contrast bare-ground and screen performance. However, the 9-wire field is always available as an alternative to a screen and may be easier to install.)</p>
|
||||
<center>
|
||||
<img src="ao12-t3.gif" width="626" height="305" border="2" vspace="5" hspace="9">
|
||||
</center>
|
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<p>Let's pause here to look at an important side question: how does ground quality affect the improvement level offer by the screen or the 9-wire field? I modeled the dipole at a height of 0.175 wavelength over bare ground and over the screen using several ground quality levels, all of which appear in <b>Table 3</b>. The worse the soil quality, the greater the improvement offered by the ground-level screen. Over very poor soil, the gain improvement is nearly 2 dB, but over very good soil, the improvement drops to only 0.2 dB. In no case of solid ground does the use of a screen seriously approach the level of a perfect ground, although at sea, one might come very close. The conclusion is that NVIS antennas over poorer grades of soil may benefit significantly from a screen or a 9-wire reflector. <b>The Inverted-V</b></p>
|
||||
<p>Testing NVIS inverted-V antennas adds another variable to our modeling efforts. Let's assume that we use a fixed height for the ends of the V. For safety, I placed the ends 0.05 wavelength (about 12.5' at 3.9 MHz) above ground. As I surveyed changing top heights, I restored the antenna to near resonance, which drew the ends in toward the center and increased the angle of the wire relative to the ground. <b>Figure 4</b> shows the bare ground and screen configurations. <b>Table 4</b> presents the test results, including the wire angle.</p>
|
||||
<center>
|
||||
<img src="ao12-4.gif" width="550" height="200" border="2" vspace="5" hspace="9">
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||||
</center>
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||||
<center>
|
||||
<img src="ao12-t4.gif" width="639" height="421" border="2" vspace="5" hspace="9">
|
||||
</center>
|
||||
<p>The inverted-V over bare ground shows much less gain than a dipole. The effective height of the entire wire is about 2/3 the distance between the lower and the upper ends. Hence, the peak gain is approximately the same as the dipole over bare ground at a height between 0.1 wavelength and 0.125 wavelength. Consistent with the dipole models, the peak gain for the inverted-V occurs at a top height of 0.175 wavelength.</p>
|
||||
<p>One advantage of an inverted-V is that the endwise beamwidth increases by from 10° to 20° relative to the dipole. In fact, the higher that we place the V, the more circular the pattern becomes. At a height of 0.225 wavelength, we find only a 23° difference between the broadside and the endwise beamwidth reports. The wire angle at this height is about 46°, a value that we cannot safely achieve at lower top heights. Obviously, the greater the wire angle relative to ground, the more omni-directional the pattern becomes.</p>
|
||||
<p>For any given top height, an inverted-V's effective height will be lower than a linear dipole at the same height. As a consequence, the inverted-V tends to benefit more from the presence of a ground-level screen (or its 9-wire substitute). The lower part of Table 4 shows a 1-dB or greater improvement in gain. As well, it shows a slight improvement in the circularity of the azimuth coverage due to a small shrinkage in the broadside beamwidth.</p>
|
||||
<p>One strategy for setting up a NVIS antenna system is to use a pair of inverted-V antennas--one for 80/75 meters, the other for 40 meters--using a common center support and a common feedpoint. If the antennas are at right angles to each other, interactions between the two sets of wires will be minimal. The limitation of such a system is that we end up with both bands using heights that are not optimal. 0.125 wavelength at 3.9 MHz is close to 0.23 wavelength at 7.2 MHz. While both heights fall within the scanned range for our test cases, one or the other may yield a pattern shape that is not ideal. We may use the same center support for both bands, but setting up separate antennas optimized for the best height and wire angle (over a reflector screen) may let us achieve near circularity of coverage or just the degree of pattern elongation that we need for the intended coverage area.</p>
|
||||
<p><b>The 1-Wavelength Loop</b></p>
|
||||
<p>An overlooked antenna for NVIS work is the 1 wavelength loop. Each side of the loop is only about half the length of a dipole for the same frequency. If we plan to supplement the antenna with a screen or other reflection means, the loop may prove to be more compact than a dipole or a V with a screen below. As well, we can nest loops for each lower HF band that we wish to cover. <b>Figure 5</b> shows the bare-ground and the screened configurations, the data for which appear in <b>Table 5</b>.</p>
|
||||
<center>
|
||||
<img src="ao12-5.gif" width="550" height="195" border="2" vspace="5" hspace="9">
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||||
</center>
|
||||
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|
||||
<img src="ao12-t5.gif" width="631" height="439" border="2" vspace="5" hspace="9">
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|
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<p>Compared to a dipole, the 1 wavelength loop provides slightly higher maximum gain levels and slightly more circular patterns. At the height of maximum gain, the loop pattern is about 24° less oval than the dipole pattern, as measured by the difference between the broadside and the endwise beamwidth values. For the loop, the broadside direction passes through the mid-side feedpoint and the midpoint of the opposite side. The endwise pattern passes through the two opposing two sides without a feedpoint. The use of a 1 wavelength by 1 wavelength screen below the loop at ground level provides slightly less added gain than it does for a dipole.</p>
|
||||
<p>In some respects, the 1 wavelength loop provides the best of the dipole and the inverted-V worlds. It has the dipole's gain and the V's nearly circular pattern. However, it does require 4 corner supports, and the feedpoint is well above the range for a good match to common 50-Ohm coaxial cable. The latter problem disappears if we add a 1/4 wavelength section of 70-75-Ohm cable.</p>
|
||||
<p><b>High-Gain NVIS Arrays</b></p>
|
||||
<p>The simple antennas that we have explored offer a balance between gain and beamwidth. Since we are dealing with nearly circular azimuth patterns with only one main lobe, the only way to increase upward or maximum gain is to decrease the beamwidth in one or both directions. Hence, high-gain arrays are not necessarily for everyone. The operator who needs to use elevation angles down to say 45° may obtain better results with one of the simple antennas. However, a central station that requires only short-distance communication may find an advantage in concentrating his or her signal upward. In fact, by using a common wire array configuration, we may obtain up to 6-dB additional gain.</p>
|
||||
<p>Perhaps the two most common NVIS arrays used to increase upward gain are the "Jamaica" and the "Shirley" array. Actually, both arrays are forms of the lazy-H facing the sky. Moreover, for raw upward gain, these arrays overlook the best of the lot: the extended (or expanded) lazy-H. Let's quickly sample all 3 antennas both with and without a screen (or multi-wire) supplement. In each case, the antenna itself will use the same AWG #14 copper wire common to the simple antennas. We feed each antenna element in phase with equal length lines to a central feedpoint. The feedpoint impedance will vary according to the element length, the spacing, and the characteristic impedance of the phasing lines. <b>Table 6</b> provides the modeled data for all 3 antennas.</p>
|
||||
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|
||||
<img src="ao12-t6.gif" width="631" height="405" border="2" vspace="5" hspace="9">
|
||||
</center>The Shirley array, shown in <b>Figure 6</b> uses 1/2 wavelength elements spaced about 0.65 wavelength apart. Some versions use folded dipole elements for presumed match to the phase lines, but that aspect of the antenna construction plays no role in establishing the basic gain and pattern data. At a height between 0.175 wavelength and 0.2 wavelength, the antenna shows a little over 4-dB gain over a dipole. The price that we pay for the gain is a very significant reduction in the broadside beamwidth (by nearly 70°), but not in the endwise beamwidth. See the lower part of <b>Figure 6</b>. The result is a highly elongated oval that favors the directions off the ends of the elements. The use of a 1 wavelength by 1 wavelength screen adds nearly a dB to the gain without altering the beamwidth values.
|
||||
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|
||||
<img src="ao12-6.gif" width="550" height="320" border="2" vspace="5" hspace="9">
|
||||
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|
||||
<p>The Jamaica array, shown in <b>Figure 7</b>, uses a standard lazy-H configuration: two 1 wavelength elements with a 1/2 wavelength space between them. Over bare ground, it improves upward gain by a full dB over the Shirley array, but the addition of a 1.5 wavelength by 1 wavelength screen adds less than a half-dB more. In both cases, the Jamaica height is the same as the Shirley height. One way to view the circularity of the patterns is to take the ratio of the broadside to endwise beamwidth values. The dipole over bare ground shows a broadside-to-endwise ratio of 1.7:1. In contrast, the Jamaica array has a ratio of less than 1.3:1 over bare ground, as illustrated by the patterns in the lower half of <b>Figure 7</b>. Both numbers are drawn from the height of maximum gain. (In contrast, the Shirley array showed a broadside-to-endwise ratio of 0.6:1.) Note that the use of collinear half wavelength elements and half wavelength spacing yields no sidelobes. The wider spacing of the Shirley elements revealed the emergence of low-angle broadside lobes.</p>
|
||||
<center>
|
||||
<img src="ao12-7.gif" width="550" height="320" border="2" vspace="5" hspace="9">
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</center>
|
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<p>For raw gain, we can do little better than increase the element length to 1.25 wavelength and use a 0.65 wavelength space between the elements. <b>Figure 8</b> shows the results, along with patterns that reveal the emergence of sidelobes in both the broadside and endwise directions. The user will have to determine whether the lower-angle sidelobes present a danger of increased noise pick-up based on the level and types of noise sources for the given location.</p>
|
||||
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|
||||
<img src="ao12-8.gif" width="550" height="320" border="2" vspace="5" hspace="9">
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</center>
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<p>If the sidelobes do not pose a problems, then the extended lazy-H array adds nearly 2 dB to the gain offered by the Jamaica beam, without a screen or with the requisite 2 wavelength by 1 wavelength screen needed by the extended lazy-H. (Of course, a field of wires about 1.5 wavelength long and extending about 0.4-0.5 wavelength beyond the broadside limits of the active array may substitute for the screen.) The screen adds only about 0.4-dB gain to the bare ground version of the antenna. Essentially, the extended lazy-H configuration provides 12.5-13 dBi maximum gain over average ground, compared to 6.4 to 7.1 dB for a dipole at roughly the same height. In exchange, as shown by the lower part of <b>Figure 8</b>, we further narrow the beamwidth--down to 44° broadside and 31° endwise. The 1.4:1 ratio shows fairly good azimuth circularity.</p>
|
||||
<p>Assuming that we can handle the Lazy-H array feedpoint impedance values with an antenna tuner, the extended version of the array offers a further unadvertised benefit that stems from its higher gain level. We can afford to place an extended lazy-H a bit higher than the optimal height and set the element length at 1.25 wavelength on 7.2 MHz, with a 40-meter spacing of about 0.6 wavelength to 0.65 wavelength. The gain deficit relative to an optimal height will be small. At 3.9 MHz, the array will be a little over half as high, and the element lengths and the spacing will be half as much. The array will still perform well, with a gain level intermediate between a dipole and a full extended lazy-H. At the lower frequency, we would find no sidelobes.</p>
|
||||
<center>
|
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<img src="ao12-9.gif" width="672" height="220" border="2" vspace="5" hspace="9">
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</center>
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<p>The lure of additional gain often blinds us to other considerations that may affect our operation. Throughout these notes, I have tried to give equal strength to gain and beamwidth comparisons. Which factor requires greater weight in deciding on a NVIS antenna requires an operator decision. If operations require more than short range, then the added gain of the lazy-H configurations may not be an advantage. Therefore, <b>Figure 9</b> may be of interest. It shows overlaid patterns for a 1 wavelength loop and for the extended lazy-H array, both over bare ground. The increased gain potential of the lazy-H captures our initial attention. However, if we count upward to the 45° elevation angle, we discover that the loop has a significantly higher gain in that elevation direction. In fact, at that angle, the lazy-H has almost no gain, since it is the angle for the null between the main lobe and the sidelobe in both the broadside and the endwise patterns.</p>
|
||||
<p><b>Conclusion</b></p>
|
||||
<p>We have looked at a variety of central-station NVIS antennas for amateur use with an eye toward finding the optimal height of maximum gain (between 0.175 wavelength and 0.2 wavelength). We also examined the level of pattern circularity achieved by these simple designs. We also explored a few high-gain NVIS arrays, as well as the level of benefit offered by ground screens or multi-wire substitutes. Which antenna might be correct for you depends on your available space, your access to supports, and also the type of operating that you do. For casual chats with a neighbor who lives beyond yon hill (or even beyond the hill beyond yon hill), a high gain array may be a nearly ideal NVIS antenna.</p>
|
||||
<p>For the emergency operator, high gain alone may not solve all challenges, especially if there is a need to reach beyond short range into the intermediate range that still falls within the skip zone. One of the simpler antennas may better serve the requirement. In his article, Straw noted the need for multiple relays to route important messages outward and then back inward toward targets, many of which required NVIS-type propagation. Under these conditions, the right antenna--abetted by high operator skill and experience--proved invaluable. In fact, it gave a contemporary rationale for continuing to call our national organization the American Radio Relay League. Indeed, an operator who copies precisely and relays accurately is as important as the antenna that he or she uses.</p>
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||||
<p><a href="ao12-models.zip">ao12-models.zip</a>.</p>
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<img src="../images/colorbar.gif" width="591" height="3">
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<p><i>Updated 01-01-2007. © L. B. Cebik, W4RNL. This item first appeared in</i> QEX<i>, Jan/Feb, 2007, pp. <b>55-61</b>. Reproduced with permission. Copyright ARRL (2007), all rights reserved. This material originally appeared in</i> QEX: Forum for Communications Experimenters <i>(<a href="https://www.arrl.org/qex" target="_blank">www.arrl.org/qex</a>).</i></p>
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<p><a href="ao.html"><img src="../images/return.gif" align="bottom" width="40" height="40"> Return to series index page</a></p>
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<p class="topics-footer">Topics: <a href='../topics.html#dipole'>Dipole</a>, <a href='../topics.html#hf-lower'>HF Lower</a>, <a href='../topics.html#horizontal-loop'>Horizontal Loop</a>, <a href='../topics.html#inverted-v'>Inverted V</a>, <a href='../topics.html#loading'>Loading</a>, <a href='../topics.html#nvis'>NVIS</a></p><br>
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<title>AO 13: NVIS Antennas for Special Needs</title>
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<h1><img src="../images/qex.gif" align="left" width="150" height="83" border="2" vspace="5" hspace="25"> <img src="../images/w4rnlwwr.gif" align="right" width="99" height="83" border="2" vspace="5" hspace="25"><i>NVIS Antennas for Special Needs</i></h1>
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<h3>L. B. Cebik, W4RNL</h3>
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<p>Our last episode dealt with basic and advanced monoband NVIS antennas. Although we found a number of differences among the antennas, most of them showed reasonably circular azimuth patterns as determined by the ratio of the broadside to endwise half-power beamwidth. Virtually all of the antennas achieved their maximum upward gain at heights between 0.175 wavelength and 0.2 wavelength. We also saw clearly the relationship between gain and beamwidth, especially when we compared antennas like the dipole, the inverted-V, and the 1 wavelength loop with more complex arrays based on the lazy-H configuration.</p>
|
||||
<p>In this follow-up set of notes, we shall look at two special needs within the overall NVIS scene: a desire for directional NVIS communications and the requirement--largely a function of newer ALE potentials--for very wide-band communications. To simplify our discussions, we shall place all antennas over bare average ground (conductivity = 0.005 S/m, relative permittivity = 13).</p>
|
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<p><b>Directional NVIS Communications</b></p>
|
||||
<p>Curiously, a few years back, I received within a fairly short time period two notes via e-mail. One correspondent wished to know if there might be an effective NVIS antenna for his coastal location, since he wished to direct as much as possible of his signal inland. The second note, a few weeks later, asked is there might be an antenna for his coastal location that would allow NVIS communications out to sea with enough front-to-back ratio to quiet signals from land-based stations in the other direction. Not only do both inquiries have an affirmative response, they both might use the same antenna.</p>
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<img src="ao13-1.gif" width="450" height="420" border="2" vspace="5" hspace="9">
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<p><b>Figure 1</b> shows two candidates composed of AWG #12 copper wire and set at 45' about 0.175 wavelength) for 3.9-MHz operation. The upper left sketch shows a conventional 2-element driver-reflector Yagi. The element spacing provides a feedpoint impedance close to 50 Ohms. As we lower the height of a directional antenna, the main forward lobe angles ever more upward. The Yagi shown has a TO angle of about 49°, with an elevation beamwidth that extends to nearly overhead. The maximum gain approaches 8.2 dBi at the TO angle, with a 7.5-dB front-to-back ratio. The radiation directly upward from the Yagi is only down about 3.5 dB or so from the maximum gain, so NIVS communications at the shortest ranges will be similar in strength to using a dipole or inverted-V. However, receiving sensitivity to the rear of the array will be well down from maximum gain. The result is directive NVIS operation, whether out to sea or inland from the sea.</p>
|
||||
<p>The way to improve the NVIS directional pattern is not with more antenna gain, but with less. The more compact Moxon rectangle--consisting of AWG #12 copper wire--has a lower maximum gain value: about 7.9 dBi. However, the TO angle is about 55°, with a smoother pattern curve from front-to-rear. Hence, the zenith angle falls within the elevation pattern's beamwidth. As well, the rectangle's configuration provides an additional 2-dB of front-to-back ratio, resulting in further quieting in the unwanted direction. We might obtain similar results from the Yagi by lowering its height somewhat.</p>
|
||||
<p>Because both antennas are horizontal directional beams, even at low heights, we have a limited horizontal beamwidth. NEC-4 reports a value of 94° for the Yagi and 120° for the Moxon rectangle. These reports derive from the azimuth patterns that we took using the TO angle as the pattern elevation angle. Since both TO angles are quite high, we must use caution in accepting the reported values.</p>
|
||||
<p>Every azimuth pattern over ground specifies an elevation angle for the pattern. Only if the elevation angle is 0° (illicit in NEC-2) will the pattern itself be circular. Azimuth-equivalent patterns in free-space may use 0° elevation and also yield a far-field tracing that is flat and circular. Every non-zero elevation angle entry in fact produces a pattern based on a conical surface, as suggested by the radically high elevation angle in <b>Figure 2</b>. Let's assume that the figure has sliced the cone from tip to base and therefore shows only the main forward lobe and not the rearward lobes. The actual half-power beamwidth on the surface of the cone is the angular distance between the points on the lobe that are 3 dB down from the peak gain value.</p>
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<img src="ao13-2.gif" width="500" height="350" border="2" vspace="5" hspace="9">
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<p>Unfortunately, limitations of software force us to project the pattern onto a flat circular plotting form. World maps suffer a similar problem when we project the features of a globe onto a flat page. The problem is distortion. With azimuth patterns, we do not notice the problem with antennas that we use for long-distance communication, because we normally us fairly low elevation angles. Hence, the cone and the plotting form are very similar. However, for NVIS operations, we are interested in high to very high angles, that is, elevation angles from 45° to 90°. At these angles, the distortion can be high, and it increases as we increase the elevation angle. The right side of <b>Figure 2</b> may seem to create only mild distortion, but the sketches show a 2:1 ratio between the angles on a flat surface and on the conical surface at the left.</p>
|
||||
<p>We may quickly calculate an approximate corrected horizontal beamwidth value using a simple equation:</p>
|
||||
<p>BWa = BWr * cos(elevation angle) or BWa = BWr * sin(theta angle)</p>
|
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<p>BWa is the actual horizontal beamwidth, BWr is the NEC report of the beamwidth, and the indicated angles are the elevation or theta angles at which we take the phi/azimuth pattern. The use of an elevation angle or a theta angle will depend on the operative convention of your antenna modeling software. NEC operates using theta angles counting from the zenith downward. We convert theta angles to elevation angle by subtracting from 90°. The equation is only a handy approximation because it does not account for the fact that the pattern has side-to-side curvature on the surface of the cone, but it provides results that are as close as we need for virtually all applications.</p>
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<p>One reason that we do not question beamwidth reports for lower angle azimuth patterns relative to the values that occur in free-space patterns is that the cosine of the elevation angle is 0.9 or higher for all such angles that are 25.8° or lower. However, the cosine of the elevation angle decreases ever more rapidly toward zero as we raise the elevation angle. The Yagi reported a beamwidth of 94° at a TO angle of 49°. The cosine of 49° is 0.646 and so the adjusted beamwidth is about 62°. The Moxon reported a beamwidth of 120° at an elevation angle of 55°. The corrected value is about 69°. Although the initial reports seemed to give the Moxon a large horizontal beamwidth advantage over the Yagi, the corrected values tell us that there is not much difference. Since wire NVIS directional antennas require aiming, correcting the high-angle azimuth beamwidth reports is essential if we are to know what coverage we can expect from it. Of course, coverage does not suddenly end beyond the beamwidth limit, but it may weaken rapidly.</p>
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<p><b>Wide-Band Terminated NVIS-ALE Antennas</b></p>
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||||
<p>Automatic Link Establishment (ALE) equipment has become almost standard in military circles. By very rapid scanning and synchronization of codes and return codes, the central station can select the most promising frequency for successful communications. Other governmental agencies have seen a potential for adapting this system to emergency communications. Since the scanning central transceiver does its work so rapidly, it does not have time for the delays associated with changing matching networks to effect a match between the equipment and the antenna. As a result, antennas that exhibit a constant impedance over a very wide frequency range have once more become very popular government acquisition items.</p>
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<p>Unfortunately, radio amateurs have also become enamored with these antennas. One of the most common misconceptions associated with these antennas is that they perform in all respects as smoothly across the frequency spectrum as the very low SWR value suggests. In general, we can rarely obtain such smooth performance over a 2:1 frequency range using an inverted-V configuration. Let's perform a small experiment. We can use the 3 antennas shown in <b>Figure 3</b> as samples. Each antenna is 42.5' high at the center and 2.5' above average ground at the ends. The distance between the center pole and the wire end is 50'. Each leg is therefore 64' long, for a total AWG #12 wire length of 128'. These dimensions apply to all three variations on the inverted-V installations. The operating range is 3.75 MHz to 7.5 MHz. Each antenna is initially a bit shorter than 1/2 wavelength electrically so that we may obtain adequate NVIS patterns throughout the passband.</p>
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<p>The single-wire unterminated V forms a performance baseline for our comparisons. At the antenna feedpoint, we shall expect (and ignore in this context) a wide set of excursions for the resistive and reactive components. The resistance will begin quite low and reach a very high value at or near the upper frequency limit. The reactance begins as a moderate capacitive value and climbs to a very high inductive value. These progressions are normal for an antenna that begins at just under 1/2 wavelength and reaches about 1 wavelength at the upper frequency limit.</p>
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<p>More significant for the present context are the data in <b>Figure 4</b>, which graphs the maximum gain and the broadside and endwise beamwidth values. The gain declines at the lower end of the spectrum as the antenna reaches 1/2 wavelength and then becomes even shorter. Otherwise, the gain level is relatively smooth and consistent with values that we saw for the inverted-V in the last episode. The broadside beamwidth shows a continuous climb with increasing frequency. In contrast, the endwise beamwidth remains relatively constant until we near the upper end of the passband.</p>
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<p>The middle sketch in <b>Figure 3</b> shows perhaps the most popular terminated wide-band antenna used by amateurs: the terminated folded dipole. There are many versions and a number of myths surrounding the antenna. The spacing between the AWG #12 conductors can vary from an inch to a foot or two with no modelable difference in the basic performance. Far more critical to any application is the length. An ideal terminated wide-band folded dipole antenna will be at least 1/2 wavelength long at the lowest operating frequency to remain above the performance "knee." The knee represents an electrical length below which performance falls off precipitously as the antenna feedpoint resistance without the termination would decrease from about 70 Ohms toward zero. The series termination would provide a stable feedpoint impedance, but also dissipate more and more of the energy. In the inverted-V configuration, the baseline non-terminated impedance would be closer to 50 Ohms. Above the knee region, the impedances undulate around the value of the terminating resistor. A resistor in the vicinity of 800 to 900 Ohms tends to yield the smoothest SWR curve. Of course, the antenna must include an effective method of converting the high terminal impedance down to a conventional coaxial-cable value.</p>
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<p>Like the single wire unterminated inverted-V, the terminated folded version is slightly short relative to the knee in order to produce acceptable patterns for NVIS operation within the 3.75- to 7.5-MHz passband. Because the amount of shortening below 1/2 wavelength is small, we should see only the start of the gain decline below about 4.5 MHz. The graph in Figure 5 confirms this suspicion. Below about 4.5 MHz, the curve takes a sharper downward direction than the corresponding gain curve for the unterminated inverted-V. Equally significant in <b>Figure 5</b> are the beamwidth curves for broadside and endwise directions from the wire antenna. Compare these curves to the corresponding set in <b>Figure 4</b>. The fundamental similarity is inescapable.</p>
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<p>An alternative method of developing a wide-band terminated antenna is to place the terminating resistors at the wire ends. These resistors require a ground-return line to provide them with a low impedance common point. The model for this basic configuration uses a wire that is 0.001 wavelength above ground so that the model will run on both NEC-2 and NEC-4.1 Commercial versions of the antenna show variations on the basic theme. Some use fans of different-length elements joined at the terminated ends. Others bring the common line back up to the feedpoint. Most of the variations tend to yield somewhat smoother SWR curves. The modeled basic version uses a 500-Ohm resistor at each end of the wire, with a 1000-Ohm reference impedance. As with the folded dipole type of terminated antenna, the system needs an effective impedance transformation device to allow a coaxial feedline. One difference between the end-terminated system and the folded-dipole type occurs with the SWR performance below the knee region. Whereas the folded version shows a smoother SWR curve, the end-terminated version tends to show a rise in SWR. Hence, the SWR provides a warning apart from gain performance to tell the user when the antenna is too short.</p>
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<p><b>Figure 6</b> graphs the maximum gain and the beamwidth values for the end-terminated antenna in its simplest form. The gain curve roughly parallels the folded-dipole curve, but with a slightly lower knee-frequency region--closer to 4 MHz than to 4.5 MHz. More interesting are the beamwidth curves. The broadside beamwidth is very close to what we found for the unterminated and the folded-dipole antenna types. However, the endwise beamwidth shows a greater variation across the passband than we found with the other two antennas. As well, the average endwise beamwidth is perhaps 25° greater and exceeds the broadside beamwidth throughout the passband. One consequence of this behavior difference is that commercial versions of the antenna tend to favor flatter installations than the 38° slope of the wires in these test models. The more level the wire, the lower the endwise beamwidth of the end-terminated wire, with a resulting equalization of beamwidths for NVIS operation.</p>
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<p>We cannot escape an ultimate comparison of the gain curves for the 3 sample antennas. See <b>Figure 7</b>. The gain differential between the two terminated antennas is inconsequential compared to the deficit relative to the unterminated wire. The gain difference runs from a minimum of 5+ dB up to over 6.5 dB across the 2:1 frequency span. Since NVIS operations often call for working at the margins of acceptable transmitted and received signals, a central station using a terminated antenna loses 3/4 of its transmitted power and over an S-unit of received signal strength compared to a simple unterminated wire. While terminated antennas might be used at field installations to simplify operation, they do not appear to offer the amateur operator any significant benefits for a durable installation.</p>
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<p>The reason for limiting the 3 antennas to a 2:1 frequency span results from the fact that terminating a given length antenna does not change the far-field pattern shape. To illustrate this basic fact, <b>Figure 8</b> presents some samples of overlaid endwise elevation patterns. Only the terminated folded-dipole antenna appears as the lower-gain entry for comparison with the unterminated antenna. Adding the end-terminated antenna would only create murky pattern outlines for the inner or weaker patterns. The essential feature of these patterns is the strict congruence between the stronger and the weaker patterns across the passband. Termination affects the radiated energy of an antenna, but not its patterns. I have included a pattern for 10 MHz to establish the reason for limiting the antenna passband. However smooth the SWR curve may be for a given terminated design, the bell-shaped pattern at 10 MHz indicates an endwise beamwidth that is usually too narrow for most NVIS operations.</p>
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<p><b>The Search for a Wide-Band Unterminated NVIS Antenna</b></p>
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<p>ALE has added a fresh incentive to find a NVIS antenna that provides full unterminated wire antenna gain but with an exceptionally wide operating bandwidth. A more ideal antenna would exhibit relatively uniform gain directly upward with equal beamwidth values, that is, with a virtually circular azimuth pattern at any elevation angle. The half-power beamwidth values should fall within a range from 70° to perhaps 110° to assure adequate energy and receiving sensitivity at about a 45°-elevation range to allow intermediate as well as short-range communications by this mode. The search for such an antenna has taken two basic directions--one relatively futile, the other more promising.</p>
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<p>Both directions share a common feature: the use of frequency-independent design techniques. The most obvious technique is to apply LPDA design equations to a dipole array that points straight upward (or downward). In general, an LPDA design with linear elements fails to achieve the desired results due to a conflict between LPDA design criteria and the height range within which NVIS antennas perform best. For any number of elements covering any frequency range, the best height falls within the 0.125 wavelength and the 0.225 wavelength range. To obtain the best LPDA performance in terms of gain and feedpoint impedance we must increase the value of one of the calculation constants (sigma) to a value higher than we find in most horizontally oriented arrays. For the 2.5-11-MHz range (a 4.4:1 frequency span), the result would be an exceptionally tall array. As well, the gain would yield beamwidths well below the desired 70° minimum value.</p>
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<p>If we shorten the array by reducing the value of sigma and reduce the number of elements, then LPDA performance over wide frequency range tends to collapse. Numerous anomalous frequencies will appear. At these frequencies, the pattern will show multiple lobes in unwanted directions and the main lobe will not be upward. As well, the impedance curve tends to vary widely. In general, the wider the frequency span for an LPDA, closer that the design constants of tau and sigma must come to the highest permissible values. As a result, if we try to place each element at a height that favors NVIS operation, we end up with an array that fails completely in its mission.</p>
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<p>An alternative array that also uses basic LPDA principles in an expanded format has become popular among some commercial antenna makers for the corporate and government market: the log-spiral doublet array. <b>Figure 9</b> shows the general outline for one such design. The spiral uses a factor of a<sup>theta</sup> to define the rate of radius increase with each wire segment. The design shown uses a pair of 4-turn spirals 180° apart to end up with the full structure.</p>
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<p>Log-spiral antennas have their greatest use in the UHF region where the design can attend to the element diameter (or strip width on a substrate). A wire version in the HF range that uses a single wire diameter will thus be less ideal and subject to numerous finicky requirements. Most commercial designs to not use an inner limiting radius for the spiral. As well most use a conical or modified conical shape (with 4 or 6 sides). Some offerings use outer-end resistive terminations, although this practice seems to defeat the point of using such a complex structure, namely, to achieve full wire gain over a wide frequency span. Hence, the state of the art relative to log-spiral NVIS potentials remains uncertain. Nevertheless, <b>Table 1</b> shows the modeled performance of the flat log-spiral design and thus suggests what may be possible some time in the future. Note that at only 2 frequencies does the beamwidth drop to less than 70° in one of the sampled planes. Patterns do not go completely askew until we pass 10.5 MHz. The sample array used here requires a 50-meter diameter just to contain the wire, with additional space needed for supports.</p>
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<p><b>Conclusion</b></p>
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<p>We began by identifying two special needs sometimes associated with NVIS operation. Obtaining a directional NVIS array proved to be the simpler project, since any number of relatively low-gain horizontal arrays might meet the operational criteria for an installation. Very-wide band NVIS operation across a large frequency spread presented the more difficult challenge. If we require relatively slow frequency changes, then we can easily press an inverted-V into service over a 2:1 frequency range for acceptable NVIS patterns. However, the higher rates of ALE frequency change require an antenna that requires no alteration of the impedance matching network in the course of operation. Some typical wide-band terminated antenna types can meet the need, with 2 restrictions. First, for many antennas that we might set up as inverted-Vs or as linear doublets, the operative frequency range for acceptable NVIS patterns remains about 2:1. Second, termination yields considerable loss of gain. Achieving a wide-band unterminated antenna design capable of covering 2.5 to 10 MHz with roughly similar gain levels and acceptable NVIS patterns without a need to switch networks remains a future goal for amateurs and an expensive project for commercial and government installations.</p>
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<p><a href="ao13-models.zip">ao13-models.zip</a>.</p>
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<p><i>Updated 05-01-2007. © L. B. Cebik, W4RNL. This item first appeared in</i> QEX<i>, May/Jun, 2007, pp. 50-55. Reproduced with permission. Copyright ARRL (2007), all rights reserved. This material originally appeared in</i> QEX: Forum for Communications Experimenters <i>(<a href="https://www.arrl.org/qex" target="_blank">www.arrl.org/qex</a>).</i></p>
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<p><a href="ao.html"><img src="../images/return.gif" align="bottom" width="40" height="40"> Return to series index page</a></p>
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<p class="topics-footer">Topics: <a href='../topics.html#doublet'>Doublet</a>, <a href='../topics.html#hf-lower'>HF Lower</a>, <a href='../topics.html#inverted-v'>Inverted V</a>, <a href='../topics.html#moxon'>Moxon</a>, <a href='../topics.html#nvis'>NVIS</a>, <a href='../topics.html#yagi'>Yagi</a></p><br>
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<title>AO 14: Reversible Wire Beams for Lower HF Use</title>
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<h1><img src="../images/qex.gif" align="left" width="150" height="83" border="2" vspace="5" hspace="25"> <img src="../images/w4rnlwwr.gif" align="right" width="99" height="83" border="2" vspace="5" hspace="25"><i>Reversible Wire Beams for Lower HF Use</i></h1>
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<h3>L. B. Cebik, W4RNL</h3>
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<p>Rotatable directional beams for the lower HF region (from 30 meters downward) tend to be rare, heavy, and expensive. Many operators choose instead to use wire beams, despite their fixed orientation. One partial way around the limitation is to build a reversible wire beam. We can accomplish this feat with varying degrees of electrical complexity by switching sections of elements or using loading components.</p>
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<p>In these notes, we shall examine 5 different but inter-related ways of achieving the goal of reversibility with the minimum of complexity. Some designs may require more real estate, but basic construction will only involve stringing and supporting elements. Any remote switching that we do will occur outside the antenna geometry. We shall look at two designs that use shorted stubs to load driver elements to become electrically long enough to serve as reflectors in 2-element beams: one Moxon rectangle and one Yagi. Then we shall look at an alternative reversible 2-element Yagi that requires 3 wires. The next step is to extend that technique to a 3-element Yagi with 5 wires. Finally, we shall reduce the wire count to 4, using an idea passed along to me by Bill Desjardins, W1ZY.</p>
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<p><b>Some Directional Beam Basics</b></p>
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<p>Although we may apply the general ideas in these notes to any of the bands in the lower HF region (or even to 160 meters), we shall focus on 40 and 30 meters, using test frequencies of 7.15 and 10.125 MHz. 30 meters is a band that is ideal for a reversible wire beam, because we can usually manage a 50-foot height. However, the band is so narrow that we do not receive a full impression of wire-beam capabilities and limitations unless we also include 40 meters.</p>
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<p>One limitation of horizontal beams is strictly frequency-related. The height of a beam determines to a significant measure its performance potential, and we measure the height in wavelengths. In the lower HF region, height is normally a fraction of a wavelength, since even at 30 meters, a wavelength is about 100'. At 40 meters, that same 100; is only about 0.7 wavelength. <b>Figure 1</b> shows selected elevation patterns at different heights from 0.25 wavelength up to 1.0 wavelength for a 2-element Yagi. At the lowest height, the beam is best used as a directional NVIS antenna. The lowest usable height for reliable DX work may be about 3/8 wavelength <b>Table 1</b> provides performance data at 0.125 wavelength increments.</p>
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<p>Note that the rise in forward gain is not smooth as we increase height. As well, the front-to-back ratio fluctuates with height, reaching its maximum value at 1/2 wavelength intervals beginning at 3/8 wavelength (with the second maximum value at about 7/8 wavelength. The value curves grow ever smoother as we exceed a 1 wavelength height, but such heights are impractical for most reversible wire beam builders.</p>
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<p>The performance that we can obtain depends on the beam that we select. Most wire antenna beam builders choose a directional beam over a bi-directional antenna to obtain freedom from rearward QRM. Different beam designs are better or worse in this department. <b>Figure 2</b> shows the elevation and azimuth patterns of 3 different beams at a height that maximizes the front-to-back ratio: 3/8 wavelength. The standard 2-element Yagi barely achieves 12 dB. The Moxon rectangle and the 3-element Yagi do far better. Similar relationships would show up at all heights, but the patterns for a height of 3/8 wavelength are simply more vivid. As we shall see, however, the front-to-back ratio will not be our only consideration if we choose to build a reversible beam.</p>
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<p>All wire beams will fail to cover a band as wide as 40 meters. <b>Figure 3</b> overlays 50-Ohm SWR patterns for the 3 subject beams. The Moxon rectangle and the 2-element Yagi have natural 50-Ohm feedpoint impedances. For greater gain in the 3-element Yagi, I selected a design with a 30-Ohm feedpoint impedance and fitted it with a Regier series matching system using 50-Ohm and 75-Ohm cables to arrive at 50 Ohms. However, the SWR curve does not significantly change its 2:1 SWR bandwidth in the conversion. All three beams manage to cover a little over half of 40 meters. Hence, for most bands (with the exception of the 50-kHz 30-meter band), the builder will have to select the preferred portion of the band. Unlike simple dipoles, the directional beams will change their gain and front-to-back properties with frequency within any of the wider bands. 2-element (reflector-driver) beams shows a descending gain value with rising frequency, while a beam with at least one director will show a rising gain value with increasing frequency. Perhaps the more critical parameter is the front-to-back ratio, which tends to go to pot for the subject beams at about the limits of the 2:1 SWR values.</p>
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<p>The final factor that limits the utility of a reversible beam is geographic. Reversible beams are useful only if there are desired communications target areas at approximate 180° bearings relative to the beam. In Tennessee, a reversible beam would cover most of Europe on one side and the VK and ZL regions in the other. Only if you are comparably situated should you consider a reversible beam.</p>
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<p><b>The Survey Elements</b></p>
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<p>Our short survey of techniques of easily reversing beams will use 4 elements. Of course, there will be text, along with a general outline of the beam under discussion. The outline graphic will also show a free-space E-plane pattern for reference and comparison apart from any particular height above ground. We shall also examine parts of two tables. <b>Table 2</b> lists the dimensions of each beam, while <b>Table 3</b> provides free-space performance data at the design frequencies.</p>
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<p>All modeled wire beam designs in <b>Table 2</b> use AWG #12 copper wire. Design techniques call for slightly different dimensions for the 30- and 40-meter Moxons, even when one measures the elements in terms of wavelengths. However, for the wire Yagis, you may use the same dimensions in wavelengths on both bands. If you design the Yagis for 40 meters, you may simply scale the element lengths and the spacing values for 30 meters. The failure to compensate for the constant element diameter only lowers the resonant frequency a bit on 30 meters, but the 50-Ohm SWR does not reach 1.5:1 on this narrow band. On 40 meters, the design frequency is 7.15 MHz. You may rescale the design upward or down ward within the band without regard to the element diameter and not incur any adverse effects.</p>
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<p>The free-space performance values in <b>Table 3</b> allow a direct comparison from one beam's potential to another's. The top section of the table provides data on the modeled performance of non-reversible forms of each beam to permit an evaluation of the performance of reversible versions. You may revise the EZNEC models for the options to place each beam at the height that you plan to use before you reach any final conclusions. The Yagi performance data include some special entries labeled "Unused Driver Open" and "Unused Driver Shorted." We shall explain those notations as we move through the various Yagi designs.</p>
|
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<p><b>The Stub-Loaded 2-Element Moxon Rectangle</b></p>
|
||||
<p>The Moxon rectangle is a 2-element driver-reflector parasitic beam that uses two forms of element coupling to arrive at the design-frequency patterns shown in <b>Figure 4</b>. Not only do we have coupling between the parallel portions of the elements, but as well between the ends of the element tails. For versions of the rectangle requiring a 50-Ohm feedpoint impedance and using a uniform element diameter throughout, a design aid is available in the form of a spreadsheet at <a href="../../content/trans/ant-design.html">Antenna Design</a>. (The spreadsheet also contains design aids for monoband quad beams and 3-element Yagis.)</p>
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<p>Ordinarily, the Moxon rectangle reflector tails are longer than shown in <b>Figure 4</b> and in <b>Table 2</b>. However, to make the beam reversible, we use two identical driver elements along with the prescribed gap between element tails. To the center of each element we connect a section of 50-W coaxial cable. At any one time, one section of cable connects to the main feedline, also a 50-Ohm cable. The other section we short to form a j65-Ohm inductive reactance. At 40 meters, the stub's electrical length is 240.4", while at 30 meters, the electrical length is 169.8". Both lengths of coax are more than long enough to reach a center point, even using a cable with a 0.66 velocity factor (and shortening the physical length accordingly). Since the cables are for independent elements, a remote DPDT switch or relay changes each coax function. It is possible to use long lines that reach the shack for these functions. By careful calculation (including the line's velocity factor), we can determine long line lengths that will serve as the required stub. In fact, we can also make the lines long and tune out the excess inductive reactance with a capacitor. This type of system would place all switching inside the warmth of the shack. However, these refinements are beyond the scope of these introductory notes.</p>
|
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<p>As shown in the performance data in <b>Table 3</b>, the one drawback of using stubs to reverse the direction of a wire Moxon rectangle is a decrease in the feedpoint impedance. The resistive impedance drops by about 8 Ohms relative to an independent version of the antenna. Squaring the rectangle a small amount by shorting dimension A and increasing the tail length, C, of both elements would raise the impedance, but at a slight cost in forward gain, plus a refiguring of the required shorted stub reactance.</p>
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<p><b>The Stub-Loaded 2-Element Yagi</b></p>
|
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<p>We may apply the same technique to a 2-element Yagi, as shown in <b>Figure 5</b> and in the tables. Because we wish to preserve the 50-Ohm impedance of the array once we install a coax section on each driver element, we increase the element spacing from 0.146 wavelength to 0.155 wavelength. The shorted reflector stub provides j75 Ohms inductive reactance. On 40 meters, the coax (electrical) length is 258.8", while on 30 meters, the length is 182.3". The two coaxial cable lengths allow an easy mid-point meeting for switching cable functions from driver to stub and back again, even allowing for the line velocity factor.</p>
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<img src="ao14-5.gif" width="475" height="200" border="2" vspace="5" hspace="9">
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<p>The switched version of the antenna slightly outperforms the independent 2-element Yagi for interesting reasons. First, the physical proportions of the elements differ. Second, a loaded reflector provides slightly better front-to-back values than a full-size reflector. Nonetheless, we would be very hard pressed to notice the difference in operation.</p>
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<p>Whether you choose a Moxon rectangle or a Yagi, the stub-loaded switched beam provides the most compact route to a reversible beam. I have seen 3-element Yagis with switched elements, normally a function swap between the director and the reflector. Often such beam use open stubs for the director to provide capacitive reactance to shorten the element's electrical length. The driver remains unswitched. Hence, each parasitic element switches between open and shorted modes, with potential length changes for each mode. Since these cables are weighty, some designs have used lumped components. In most cases, the switch systems are more complex than the ones used for the 2-element beams. As well, few Yagi 3-element designs provide peak performance with equal spacing between each pair of elements, and very often they also require a matching network to raise a low feedpoint impedance (perhaps 25 Ohms) to the main feedline value. We may keep our switching simple if we have enough land to add some further wire elements.</p>
|
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<p><b>The 3-Wire 2-Element Yagi</b></p>
|
||||
<p>One way to simplify switching is to begin with a common reflector element. For 2-element Yagis, we then add a driver on each side of the reflector. As shown in <b>Table 2</b>, the spacing of the elements returns to the value of the independent beam. The net result, as shown in <b>Figure 6</b>, is essentially two 2-element Yagis with a common reflector. Instead of switching loading stub lines, this design simply switches the feedline. We can handle this task at the antenna level or we can bring the two feedlines to the shack and use a simple mechanical switch.</p>
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<img src="ao14-6.gif" width="450" height="205" border="2" vspace="5" hspace="9">
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<p>The 2-element reflector-driver Yagi has a limitation. As shown by the modeled data in <b>Table 3</b>, the performance changes according to whether the unused driver is shorted across the feedpoint or left open. The open connection provides the better front-to-back ratio. (The free-space E-plane pattern in <b>Figure 6</b> shows the results of an open unused driver.) When the front-to-back ratio depends wholly on a reflector element, the unused element exerts a greater effect on the reflector performance (that is, on the current magnitude and phase angle), than when we have a parasitic beam with at least one director.</p>
|
||||
<p>Obtaining an open condition in the unused driver is a simple matter electrically if we switch at the antenna level. However, the arrangement may prove to be mechanically complex. We may also achieve the desired condition with a switch at the shack end of the feedlines if we attend to the cable lengths. If the feedlines are an odd multiple of a quarter wavelength, then shorting the unused line will produce an open circuit at the feedpoint. An even multiple of a quarter wavelength will require an open line end to yield an open circuit at the feedpoint. (In calculating the line lengths, of course, be sure to include the line's velocity factor.)</p>
|
||||
<p><b>The 5-Wire 3-Element Yagi</b></p>
|
||||
<p>The 3-wire reversible beam for 2-element Yagis becomes a 5-wire reversible beam for 3-element Yagis. Such an array requires considerable open ground, as the dimensions in Table 2 and the outline in <b>Figure 7</b> reveal. In the process, we gain an advantage but acquire a disadvantage.</p>
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<img src="ao14-7.gif" width="520" height="225" border="2" vspace="5" hspace="9">
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<p>The "bad" news first: to make the effort of installing a 3-element wire beam worthwhile, we need to select a design that provides significant gain over the 2-element Yagis. The data in <b>Table 3</b> show that the design provides 1.5 to 2 dB additional forward gain over either 2-element beam. In the process, the feedpoint impedance drops to about 30 Ohms. Therefore, we require a matching system to reach 50 Ohms in order to achieve the widest possible operating bandwidth. We can easily achieve the match using common 50-Ohm and 75-Ohm coaxial cable sections in a Regier series match. We shall not go into the required calculations for this type of match in this set of notes. However, you can download a spreadsheet that includes the Regier series match within a collection of common matching systems at <a href="../trans/ant-match.html">Antenna Matching</a>. (The collection includes 3 types of series matches, the beta match, 2 versions of gamma-match calculations, and the matchline-and-stub system. Like the antenna design aid, the spreadsheet appears in both Quattro Pro and Excel formats.) The series matching system transfers matching complexities from the feedpoint junction to the transmission line itself. Therefore, it adds no further weight to the system.</p>
|
||||
<p>The "good" news is that the front-to-back ratio of a 3-element Yagi is largely a function of the director and not the reflector element. Therefore, as is clear from the data in <b>Table 3</b>, the state of the unused feedpoint makes almost no difference to the array performance in either direction. With a series match in place, it is likely that switching at the shack may be the easiest way to handle directional changes for the array.</p>
|
||||
<p><b>The 4-Wire 3-Element Yagi</b></p>
|
||||
<p>W1ZY determined that he did not have the real estate needed for the 5-wire beam. He also reasoned that he could feed the driver with parallel transmission line and use his antenna tuner to arrive at the impedance needed by his equipment. Therefore, he opted for the scheme shown in <b>Figure 8</b>. As the free-space E-plane pattern suggests, he lost no performance in his truncated 4-wire version of the reversible 3-element Yagi.</p>
|
||||
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<img src="ao14-8.gif" width="490" height="245" border="2" vspace="5" hspace="9">
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<p>The dimensions in <b>Table 2</b> reveal that the elements consist of 2 directors and 2 "reflectors." I place the word reflector in quotation marks, because each of these two elements trades functions as we swap directions. One of the two elements becomes a reflector. The element between it and a director becomes the driver. The driver length makes little difference to beam performance. (In fact, J-poles have been used as 3-element Yagi drivers.) Driver length becomes critical only if we are seeking a particular feedpoint impedance. In this design for a reversible beam, parallel feedline allows the use of a driver impedance with a considerable reactive component. Therefore, we may do the work of 5 elements with only 4 and still obtain full 3-element Yagi performance in each direction.</p>
|
||||
<p>For optimal performance, we must observe some cautions with the 4-wire reversible beam. First, in order to function correctly as a reflector, the unused driver must be shorted across the feedpoint by a relay or by a precisely cut line. With the reflector open at the center, the beam loses almost a full dB of forward gain and most of the high front-to-back ratio shown in the free-space E-plane pattern in <b>Figure 8</b>.</p>
|
||||
<p>Second, as the SWR increases at the feedpoint, even low-loss parallel lines begin to show detectable losses. The values at the feedpoint (about 35 + j50 Ohms) are not fatal. But losses over a 100' length of line may exceed 0.5 dB. As well, the values that appear at the tuner terminals may exceed the matching range. At 40 meters, the use of one of the modern low-loss coax cables may prove to match parallel line loss with an easier match at the tuner. In addition, the low-loss coax lines need no spacing between each other to prevent unwanted coupling. Nevertheless, in both cases, the lines require cutting to a length that-with the switching system used-will ensure a closed circuit at the active reflector (inactive driver) center.</p>
|
||||
<p><b>Conclusion</b></p>
|
||||
<p>We have looked at 5 of many variations on the theme of creating a reversible horizontal wire beam for the lower HF region. The options that we explored are among the simplest electrically, although some of them required a considerable open area for implementation. The chief goal of these notes has been to show that for the budget-minder operator with a penchant for using wire, a reversible beam is not only possible, but both feasible and practical.</p>
|
||||
<p>However much we tried to simplify the designs, we could not eliminate all electrical complexities. A reversible beam requires a switch somewhere along the line to change directions. As well, the unused element or the loading stub require attention to both line length and the switched condition at the line end. These matter are generally considerations that we wrestle with during installation. When operating, we may change directions as quickly as we can flip a switch.</p>
|
||||
<p>There are numerous variations on the design shown as examples. Perhaps the most common tendency would be to think of the elements in the form of inverted-Vs. This option has two constraints and one advantage. The advantage is the ability to provide a strong support along the centerline for the cable hanging from the elements. One of the constraints is the necessity to redesign the elements to suit the new configuration. To base the design on a few guesses derived from the linear designs shown here or elsewhere seems a bit careless when good design tools are readily available.</p>
|
||||
<p>The second constraint concerns performance. Inverted-V elements will reduce the front-to-side ratio of the beam patterns. In addition, V elements will also lower the effective height of the antenna. In the lower HF region, we are already height challenged with respect to obtaining a low-enough elevation angle for superior DX performance. At a certain (unspecified) point, lowering the ends of a horizontal beam may elevate the TO angle too much. In such cases, one might wish to consider one of the many directional vertical arrays, especially below 40 meters. Reversible beams are both possible and practical, but they will not overcome the limits that we briefly examined at the beginning of these notes.</p>
|
||||
<p><a href="ao14-models.zip">ao14-models.zip</a>.</p>
|
||||
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<img src="../images/colorbar.gif" width="591" height="3">
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<p><i>Updated 07-07-2007. © L. B. Cebik, W4RNL. This item first appeared in</i> QEX<i>, Jul/Aug, 2007, pp. 52-57. Reproduced with permission. Copyright ARRL (2007), all rights reserved. This material originally appeared in</i> QEX: Forum for Communications Experimenters <i>(<a href="https://www.arrl.org/qex" target="_blank">www.arrl.org/qex</a>).</i></p>
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<p><a href="ao.html"><img src="../images/return.gif" align="bottom" width="40" height="40"> Return to series index page</a></p>
|
||||
<p class="topics-footer">Topics: <a href='../topics.html#hf-lower'>HF Lower</a>, <a href='../topics.html#loading'>Loading</a>, <a href='../topics.html#moxon'>Moxon</a>, <a href='../topics.html#reversible'>Reversible</a>, <a href='../topics.html#yagi'>Yagi</a></p><br>
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